8.1.1 Direct Digital FSK Modulation

In fractional mode, the finest delta frequency difference between two programmable output frequencies is equal to

In other words, when the fractional numerator is incremented by 1 (one step), the output frequency will change by Δf_min. A two steps increment will therefore change the frequency by 2 * Δf_min.

In FSK operation, the instantaneous carrier frequency is kept changing among some pre-defined frequencies. In general, the instantaneous carrier frequency is defined as a certain frequency deviation from the nominal carrier frequency. The frequency deviation could be positive and negative.

Figure 51. General FSK Definition

Figure 52. Typical 4FSK Definition

The following equations define the number of steps required for the desired frequency deviation with respect to the nominal carrier frequency output at the RFoutTx or RFoutRx port.

Table 32. FSK Step Equations

POLARITY	SYNTHESIZER MODE	PLL MODE
POSITIVE SWING	Equation 4.	Equation 5.
NEGATIVE SWING	Equation 6. 2's complement of Equation 4	Equation 7. 2's complement of Equation 5

In FSK PIN mode and FSK SPI mdoe, register R25-32 and R9-16 are used to store the desired FSK frequency deviations in term of the number of step as defined in the above equations. The order of the registers, 0 to 7, depends on the application system. A typical 4FSK definition is shown in Figure 52. In this case, FSK_DEV0_Fx and FSK_DEV1_Fx shall be calculated using Equation 4 or Equation 5 while FSK_DEV2_Fx and FSK_DEV3_Fx shall be calculated using Equation 6 or Equation 7.

For example, if FSK PIN mode is enabled in F1 to support 4FSK modulation, set
FSK_MODE_SEL1 = 0
FSK_MODE_SEL0 = 0
FSK_LEVEL = 2
FSK_EN_F1 = 1

Table 33. FSK PIN Mode Example

RAW FSK DATA STREAM INPUT	EQUIVALENT SYMBOL INPUT	REGISTER SELECTED	RF OUTPUT
	10	FSK_DEV2_F1
	11	FSK_DEV3_F1
	10	FSK_DEV2_F1
	11	FSK_DEV3_F1
	01	FSK_DEV1_F1
	00	FSK_DEV0_F1
	...	...

FSK SPI mode assumes the user knows which symbol to send; user can directly write to register R34, FSK_DEV_SEL to select the desired frequency deviation.

For example, to enable the device to support 4FSK modulation at F1 using FSK SPI mode, set
FSK_MODE_SEL1 = 0
FSK_MODE_SEL0 = 1
FSK_LEVEL = 2
FSK_EN_F1 = 1

Both the FSK PIN mode and FSK SPI mode support up to 8 levels of FSK. To support an arbitrary-level FSK, use FSK SPI FAST mode or FSK I2S mode. Constructing pulse-shaping FSK modulation by over-sampling the FSK modulation waveform is one of the use cases of these modes.

Analog-FM modulation can also be produced in these modes. For example, with a 1-kHz sine wave modulation signal with peak frequency deviation of ±2 kHz, the signal can be over-sampled, say 10 times. Each sample point corresponding to a scaled frequency deviation.

Figure 53. Over-Sampling Modulation Signal

In FSK SPI FAST mode, write the desired FSK steps directly to register R33, FSK_DEV_SPI_FAST. To enable this mode, set
FSK_MODE_SEL1 = 1
FSK_MODE_SEL0 = 1
FSK_EN_F1 = 1

In FSK I2S mode, clock in the desired binary format FSK steps in the FSK_D1 pin.

Figure 54. FSK I2S Mode Example

To enable FSK I2S mode, set
FSK_MODE_SEL1 = 1
FSK_MODE_SEL0 = 0
FSK_EN_F1 =1

8.1.2 Frequency and Output Port Switching with TrCtl Pin

Register R0, RXTX_CTRL, and RXTX_POL are used to define the output port switching behavior with the TrCtl pin. To enable switching with TrCtl pin, set RXTX_CTRL=1.

Register R0, F1F2_CTRL, and F1F2_SEL define the operation of the frequency switching between the two pre-defined frequencies F1 and F2. To switch frequency using the TrCtl pin, set F1F2_CTRL to 1. F1F2_SEL selects the output frequency for the current status. For example, if the current active output frequency is F1, toggling TrCtl pin will change the output frequency to F2. Toggling TrCtl pin again will change the output frequency back to F1.

8.1.3 OSCin Configuration

OSCin supports single-end clock, differential clock as well as crystal. Register R34 defines OSCin configuration.

Table 37. OSCin Configuration

OSCin TYPE	SINGLE-ENDED CLOCK	DIFFERENTIAL CLOCK	CRYSTAL
Connection Diagram
Register Setting	IPBUF_SE_DIFF_SEL = 0	IPBUF_SE_DIFF_SEL = 1 IPBUFDIFF_TERM = 1	XTAL_EN = 1 XTAL_PWRCTRL = Crystal dependent

8.1.4 Register R0 F1F2_INIT, F1F2_MODE usage

These register bits are used to define the calibration behavior. Correct setting is important to ensure that every F1-F2 switching time is optimized. Figure 56 illustrates the usage of these register bits.

Figure 56. F1F2_INIT, F1F2_MODE Usage

Before t₀: Device initialization

• Power up the device.
• Write all registers to the device.
  • Ensure FCAL_EN = 1 to enable calibration.
  • Set F1F2_MODE = 1 to make both F1 and F2 being calibrated during initialization. If F1F2_MODE = 0, only the output frequency (F1 in this example) will be calibrated, F2 will not be calibrated. Furthermore, if F1F2 switching is triggered by the TrCtl pin, F1F2_MODE must be equal to 1.
  • Set F1F2_INIT = 0. Although the setting of this bit is irrelevant and not important here but if F1F2_INIT = 1, change it back to zero before attempting to change the frequency from F1 to F2.

At t₀: Locked to F1
After initialization, both F1 and F2 are calibrated. The calibration data is stored in the internal memory.

At t₁: Switch to F2.
Since FCAL_EN = 1, calibration will start over again when the output is switching from F1 to F2. F2 calibration begins based on the last calibration data, which is the calibration data obtained at t₀. If the environment (for example, temperature) does not change much, the new calibration data will be similar to the old data. As a result, the calibration time is minimal and therefore, the switching time will be short.

At t₂: Switch back to F1
Again, F1 calibration starts over and begins with the last calibration data as obtained at t₀. Calibration time is again very short, as is the switching time.

At t₃: Switch again to F2
This time, the calibration begins with the calibration data obtained at t₁, which is the last calibration data.

At t₄: Switch back to F1
Calibration begins with the calibration data obtained at t₂, which is the last calibration data.

At t₅: Set new F1, F2 frequency

• Write to the relevant registers to set the new F1 and F2 frequency (for example, change the N-divider values)
• Initiate calibration by re-writing register R0
  • Set F1F2_INIT=1. Both F1' and F2' will be calibrated

At t₆: Locked to F1'
F1' and F2' calibration completed and their calibration data are ready.

At t₇: Release F1F2_INIT bit
This bit has to be reset to zero or otherwise both F1' and F2' will be calibrated every time they are toggling.

At t₈: F1' calibration data is updated
Since F1F2_INIT is located in register R0, when writing F1F2_INIT = 0 to the device, calibration is once again triggered. However, only F1' will be re-calibrated, the calibration data of F2' remains unchanged.

At t₉: Switch to F2'
F2' calibration begins with the calibration data obtained at t₆, which is the last calibration data. Calibration time is again very short, as is the switching time.

At t₁₀: Switch back to F1'
F1' calibration starts over and begins with the last calibration data as obtained at t₈.

At t₁₁: Switch again to F2'
The calibration begins with the calibration data obtained at t₉, which is the last calibration data.

As illustrated above, register F1F2_INIT must be used properly in order to ensure that every F1-F2 switching time is optimized.

8.1.5 FastLock with External VCO

Fastlock may be required in PLL mode where an external VCO with a narrow loop bandwidth is desired. The LMX2571 adopts a new FastLock approach to support the very fast switching time requirement in PLL mode.

There are two control pins in the chip, FLout1 and FLout2. Each pin is used to control a SPST analog switch, S1 and S2. The loop filter value with or without FastLock is the same, except that with FastLock, one more C2 and two SPST switches are needed.

Figure 57. FastLock with SPST Switches

When LMX2571 is locked to F1, FLout1 will close the switch S1. When LMX2571 is locked to F2, either by toggling the TrCtl pin or program register R0, F1F2_SEL, S1 will be released while S2 will be closed by FLout2. Although S1 is released, the charge stored in C2a remains unchanged. Thus, when the output is switched back to F1, the Vtune voltage is almost correct, no (or little) charging or discharging to C2a is required which speeds up the switching time. For example, if Vtune for F1 and F2 are 1 V and 2 V, respectively, without FastLock, when the switching frequency shifts from F1 to F2, C2 will have to be re-charged from 1 V to 2 V — this is a big voltage jump. With FastLock, when S2 is closed, Vtune is almost equal to 2 V because C2b maintains the charge. Only a tiny voltage jump (re-charge) is required to make it reach the final Vtune voltage.

Figure 58 and Figure 59 compare the frequency switching time using different switching methods. In both cases, the loop bandwidth is 4 kHz while f_PD is 28 MHz. Figure 58 shows the switching time for a frequency jump from 430 MHz to 480 MHz with SPST switches. Frequency switching is toggled by the TrCtl pin. Switching time is approximately 1 ms. Frequency switching in Figure 59 is done in the traditional way. That is, change the output frequency by writing to the relevant registers such as N-divider values. In this case, because f_PD is very much bigger than the loop bandwidth, cycle slipping jeopardizes the switching time to more than 20 ms.

Figure 58. F1F2 Switching With SPST Switches

Figure 59. Change F1 Frequency Via SPI Programming

8.1.6 OSCin Slew Rate

A phase-lock loop consists of a clean reference clock, a PLL, and a VCO. Each of these contributes to the total phase noise. The LMX2571 is a high-performance PLL with integrated VCO. Both PLL noise and VCO noise are very good. Typical PLL 1/f noise and noise floor are –124 dBc/Hz and –231 dBc/Hz, respectively. To get the best possible phase-noise performance from the device the quality of the reference clock is very important because it may add noise to the loop. First of all, the phase noise of the reference clock must be good so that the final performance of the system is not degraded. Furthermore, using reference clock with a rather high slew rate (such as a square wave) is highly preferred. Driving the device input with a lower slew rate clock will degrade the device phase noise.

For a given frequency, a sine wave clock has the slowest slew rate, especially when the frequency is low. A CMOS clock or differential clock have much faster slew rates and are recommended. Figure 60 shows a phase-noise comparison with different types of reference clocks. Output frequency is 480 MHz while the input clock frequency is 26 MHz. As one can see there is a 5-dB difference in phase noise when using a clipped sine wave TCXO compared to a differential LVPECL clock. The internal crystal oscillator of the LMX2571 performance is also very good. If temperature compensation is not required, use crystal as the reference clock is a very good price-performance option.

Figure 60. Phase Noise vs Input Clock

8.1.7 RF Output Buffer Power Control

Registers OUTBUF_TX_PWR_Fx and OUTBUF_RX_PWR_Fx are used to set the output power at the RFoutTx and RFoutRx ports. Figure 61 shows a typical output power vs power control bit plot in synthesizer mode. VCO frequency was 4800 MHz, and channel dividers were set to produce the shown output frequencies.

Figure 61. Configurable RF Output Power

8.1.8 RF Output Buffer Type

Registers R35, OUTBUF_TX_TYPE, OUTBUF_RX_TYPE are used to configure the RF output buffer type between open drain and push-pull. Push-pull is easy to use; all that is required is a DC-blocking capacitor at the output. The output waveform is square wave and therefore, harmonics rich. Open-drain output provides an option to reduce the harmonics using an LC resonant pullup network at its output. Table 38 summarizes an example an open-drain vs push-pull application.

Table 38. RF Output Buffer Type

BUFFER TYPE	OPEN DRAIN			PUSH-PULL
Connection Diagram
Output Power	470 MHz	480 MHz	490 MHz	470 MHz	480 MHz	490 MHz
fo	2.7 dBm	2.8 dBm	2.8 dBm	–0.1 dBm	0 dBm	0.1 dBm
2fo	–31 dBc	–30.7 dBc	–30.5 dBc	–30.4 dBc	–30.2 dBc	–30 dBc
3fo	–17.3 dBc	–17.9 dBc	–18.1 dBc	–11.9 dBc	–12.1 dBc	–12.4 dBc
4fo	–39 dBc	–40.4 dBc	–41.6 dBc	–28.5 dBc	–28.4 dBc	–28.1 dBc
5fo	–18.1 dBc	–17.8 dBc	–17.6 dBc	–15.6 dBc	–15.6 dBc	–15.7 dBc
6fo	–27.6 dBc	–27.2 dBc	–28.5 dBc	–29.5 dBc	–29.8 dBc	–29.3 dBc

Clearly, with a proper LC pull up in open drain architecture, the 3^rd to 5^th harmonics could be reduced.

8.1.9 MULT Multiplier

The main purpose of the multiplier, MULT, in the R–divider is to push the in-band fractional spurs far away from the carrier such that the spurs could be filtered out by the loop filter. In a fractional engine, the fractional spurs appear at a multiple of f_PD * N_frac. In cases where both f_PD and N_frac are small, the fractional spurs will appear very close to the carrier. These kinds of spurs are called in-band spurs.

In Case I, the VCO frequency is an integer multiple of the f_PD, so N_frac is zero and there are no spurs. However, in Case II, the spur appears at an offset of 200 kHz. If this spur cannot be reduced by other typical spur-reduction techniques such as dithering, user can enable the MULT to overcome this problem. If the MULT is enabled as depicted in Case III, the spurs can be pushed to an offset of 5 MHz. In this case, the MULT together with the Post-divider changes the phase detector to a little bit higher frequency. As a consequence, the spurs are pushed further away from the carrier and are reduced more by the loop filter.

Another use case of MULT is to make higher phase-detector frequency. For example, if OSCin is 20 MHz, user can set MULT to 5 to make f_PD go to 100 MHz. As a result, the N-divider value will be reduced by 5 times; therefore, the PLL phase noise is reduced. A wide loop bandwidth can then be used to reduce the VCO noise. Consequently, the synthesizer close-in phase noise would be very good.

The MULT multiplier is an active device in nature, whenever it is enabled, it will add noise to the loop. For best phase noise performance, it is recommended to set MULT not greater than 6.

To use the MULT, beware of the restriction as indicated in the Electrical Characteristics table and Table 16.

8.1.10 Integrated VCO

The integrated VCO is composed of 3 VCO cores. The approximate frequency ranges for the three VCO cores with their gains is as follows:

8.2.1 Synthesizer Duplex Mode

In this example, the internal VCO is being used. The PLL will be put in fractional mode to support 4FSK direct digital modulation using FSK PIN mode. Both frequency (F1, F2) switching as well as RF output port switching is toggled by the TrCtl pin. MULT multiplier in the R-divider will be used to reduce spurs.

Figure 62. Typical Synthesizer Duplex Mode Application Schematic

8.2.1.2 Detailed Design Procedure

First of all, calculate all the frequencies in each functional block.

Figure 63. F1 Frequency Plan

Assign F1 frequency to be 902 MHz. With CHDIV1 = 5 and CHDIV2 = 1, the total division is 5. As a result, the VCO frequency will be 902 * 5 = 4510 MHz, which is within the VCO tuning range.

OSCin is 26 MHz, put Pre-divider = 1 to meet the MULT input frequency range requirement.

To meet the maximum MULT output frequency requirement, possible MULT values are 3 to 5. Play around the allowable MULT values and Post-divider values to get the optimum phase noise and spurs performance. Assuming MULT = 4 and Post-divider = 1 returns the best performance, then f_PD = 104 MHz.

N-divider = 21.68269231, that means N_integer = 21 while N_frac = 0.68269231. To use the direct digital modulation feature, put fractional denominator, DEN = 0. The actual DEN value is, in fact, equal to 2²⁴ = 16777216. So the fractional numerator, NUM, is equal to N_frac * DEN = 11453676.

Use Equation 4 and Equation 6 to calculate the required FSK steps. For +10 kHz frequency deviation, the FSK step value is equal to [10000 * 16777216 / (104 * 10⁶)] * (5 * 1 / 2) = 4033. For –10 kHz frequency deviation, the FSK step value is equal to 2's complement of 4033 = 61502. Similarly, the FSK step values for ±30 kHz frequency deviation are 12099 and 53436.

All the required configuration values for F2, 928 MHz can be calculated in the similar fashion and are summarized as follows:

Table 41. Frequency Plan Summary

CONFIGURATION PARAMETER	F1 (902 MHz)	F2 (928 MHz)
Pre-divider	1	1
MULT	4	4
Post-divider	1	1
PDF	104 MHz	104 MHz
VCO	4510 MHz	4640 MHz
N-divider	21.68269231	22.30769231
Ninteger	21	22
DEN	0	0
NUM	11453676	5162220
CHDIV1	5	5
CHDIV2	1	1
FSK_DEV0	4033
FSK_DEV1	12099
FSK_DEV2	61502
FSK_DEV3	53436

Assume here that the base charge pump current = 1250 µA, CP Gain = 1x and 3^rd order Delta Sigma Modulator without dithering is adopted in both frequency sets. The register settings are summarized as follows:

Table 42. Register Settings Summary

CONFIGURATION PARAMETERS	REGISTER BIT	COMMON SETTING	F1 SPECIFIC SETTING	F2 SPECIFIC SETTING
VCO calibration	FCAL_EN	1 = Enabled
Lock detect	SDO_LE_SEL	1 = Lock detect output
	LD_EN	1 = Enabled
OSCin buffer type	IPBUF_SE_DIFF_SEL	0 = SE input buffer
Dithering	DITHERING	0 = Disabled
Charge pump gain	CP_GAIN	1 = 1x
Base charge pump current	CP_IUP	8 = 1250 µA
	CP_IDN	8 = 1250 µA
MULT settling time	MULT_WAIT	520 = 20 µs
Output buffer type	OUTBUF_RX_TYPE	1 = Push pull
	OUTBUF_TX_TYPE	1 = Push pull
Output buffer auto mute	OUTBUF_AUTOMUTE	0 = Disabled
TrCtl pin polarity	RXTX_POL	0 = Active LOW = TX
TX RX switching mode	RXTX_CTRL	1 = TrCtl pin control
Enable F1 F2 initialization	F1F2_MODE	1 = Enabled
F1 F2 switching mode	F1F2_CTRL	1 = Control by TrCtl pin
Pre-divider	PLL_R_PRE_F1		1
	PLL_R_PRE_F2			1
MULT multiplier	MULT_F1		4
	MULT_F2			4
Post-divider	PLL_R_F1		1
	PLL_R_F2			1
ΔΣ modulator order	FRAC_ORDER_F1		3 = 3rd order
	FRAC_ORDER_F2			3 = 3rd order
PFD delay	PFD_DELAY_F1		5 = 8 clock cycles
	PFD_DELAY_F2			5 = 8 clock cycles
CHDIV1 divider	CHDIV1_F1		1 = Divide by 5
	CHDIV1_F2			1 = Divide by 5
CHDIV2 divider	CHDIV2_F1		0 = Divide by 1
	CHDIV2_F2			0 = Divide by 1
Internal 3rd pole loop filter	LF_R3_F1		4 = 800 Ω
	LF_R3_F2			4 = 800 Ω
Internal 4th pole loop filter	LF_R4_F1		4 = 800 Ω
	LF_R4_F2			4 = 800 Ω
Output port selection	OUTBUF_TX_EN_F1		1 = TX port enabled
	OUTBUF_RX_EN_F2			1 = RX port enabled
Output power control	OUTBUF_TX_PWR_F1		6
	OUTBUF_RX_PWR_F2			6
FSK mode	FSK_MODE_SEL1 FSK_MODE_SEL0	00 = FSK PIN mode
FSK level	FSK_LEVEL	2 = 4FSK
Enable FSK modulation	FSK_EN_F1		1 = Enabled
FSK deviation at 00	FSK_DEV0_F1		4033 = +10 kHz
FSK deviation at 01	FSK_DEV1_F1		12099 = +30 kHz
FSK deviation at 10	FSK_DEV2_F1		61502 = -10 kHz
FSK deviation at 11	FSK_DEV3_F1		53436 = -30 kHz
Fractional denominator	PLL_DEN_F1[23:16]		0
	PLL_DEN_F1[15:0]		0
	PLL_DEN_F2[23:16]			0
	PLL_DEN_F2[15:0]			0
Fractional numerator	PLL_NUM_F1[23:16]		174
	PLL_NUM_F1[15:0]		50412
	PLL_NUM_F2[23:16]			78
	PLL_NUM_F2[15:0]			50412
Ninteger	PLL_N_F1		21
	PLL_N_F2			22
Prescaler	PLL_N_PRE_F1		0 = Divide by 2
	PLL_N_PRE_F2			0 = Divide by 2

8.2.1.3 Synthesizer Duplex Mode Application Curves

Figure 64. F1 (TX) Phase Noise and Spurs

Figure 66. F1 (TX) to F2 (RX) Switching

Figure 68. F1 to F2 Switching Time

Figure 70. 4FSK Modulation

Figure 65. F2 (RX) Phase Noise and Spurs

Figure 67. F2 (RX) to F1 (TX) Switching

Figure 69. F2 to F1 Switching Time

Figure 71. 4FSK Modulation Quality

8.2.2 PLL Duplex Mode

In this example, the internal VCO will be bypassed, and the device is used to lock to an external VCO. TI’s dual SPST analog switch, TS5A21366 is used to facilitate FastLock between two frequencies.

Figure 72. Typical PLL Duplex Mode Application Schematic

8.2.2.2 Detailed Design Procedure

Again, we need to figure out all the frequencies in each functional block first.

Figure 73. Frequency Plan in PLL Duplex Mode

Follow the previous example to determine all the necessary configurations. Table 43 is the summary in this example.

Table 43. PLL Duplex Mode Frequency Plan Summary

CONFIGURATION PARAMETER	F1 (430 MHz)	F2 (480 MHz)
Pre-divider	1	1
MULT	5	5
Post-divider	3	3
PDF	28 MHz	28 MHz
VCO	430 MHz	480 MHz
N-divider	15.35714286	17.14285714
Ninteger	15	17
DEN	1234567	1234567
NUM	440917	176367

To enable external VCO operation, set the following bits:

Table 44. PLL Duplex Mode Register Settings Summary

CONFIGURATION PARAMETER	REGISTER BITS	SETTING
Charge pump polarity	EXTVCO_CP_POL	0 = Positive
External VCO charge pump gain	EXTVCO_CP_GAIN	1 = 1x
Base charge pump current	EXTVCO_CP_IUP	8 = 1250 µA
	EXTVCO_CP_IDN	8 = 1250 µA
Select PLL mode operation	EXTVCO_SEL_F1, EXTVCO_SEL_F2	1 = External VCO
CHDIV3 divider	EXTVCO_CHDIV_F1, EXTVCO_CHDIV_F2	0 = Bypass

Make sure that register R0, FCAL_EN is set so that FastLock is enabled.

The loop bandwidth had been design to be around 4 kHz, while phase margin is about 40 degrees.

8.2.2.3 PLL Duplex Mode Application Curves

Figure 74. F1 to F2 Switching

Figure 76. F1 to F2 Switching Time

Figure 75. F2 to F1 Switching

Figure 77. F2 to F1 Switching Time

8.2.3 Synthesizer/PLL Duplex Mode

This example will demonstrate the device's capability in switching two frequencies using internal and external VCO. VCO switching is toggled by the TrCtl pin. Direct digital FSK modulation is enabled in TX using FSK I2S mode.

Figure 78. Typical Synthesizer/PLL Duplex Mode Application Schematic

8.2.3.2 Detailed Design Procedure

Frequency plans in TX and RX paths are as follows:

Figure 79. TX and RX Frequency Plans

Follow the previous examples to determine all the necessary configurations. To enable FSK I2S mode, set

FSK_MODE_SEL1=1
FSK_MODE_SEL=0
FSK_EN_F2=1

8.2.3.3 Synthesizer/PLL Duplex Mode Application Curves

Figure 80. External VCO to Internal VCO Switching

Figure 82. External VCO to Internal VCO Switching Time

Figure 84. Simulated FM Modulation (10 times over-sampling)

Figure 81. Internal VCO to External VCO Switching

Figure 83. Internal VCO to External VCO Switching Time

Figure 85. Simulated FM Modulation (20 times over-sampling)