The single-channel OPA1677, dual-channel OPA1678, and quad-channel OPA1679 (OPA167x) op amps offer higher system-level performance over legacy op amps commonly used in audio circuitry.
The OPA167x amplifiers achieve a low 4.5-nV/√Hz noise density and low distortion of 0.0001% at 1 kHz, which improves audio signal fidelity. These devices also offer rail-to-rail output swing to within 800 mV with a 2-kΩ load, which increases headroom and maximizes dynamic range.
To accommodate the power-supply constraints of many types of audio products, the OPA167x operate over a very-wide supply range of ±2.25 V to ±18 V (or 4.5 V to 36 V) on only 2 mA of supply current. These op amps are unity-gain stable and have excellent dynamic behavior over a wide range of load conditions, allowing the OPA167x to be used in many audio circuits.
The OPA167x amplifiers use completely independent internal circuitry for lowest crosstalk and freedom from interactions between channels, even when overdriven or overloaded.
PART NUMBER | CHANNELS | PACKAGE(1) |
---|---|---|
OPA1677 | Single | SOIC (8) |
SOT-23 (5) | ||
OPA1678 | Dual | SOIC (8) |
VSSOP (8) | ||
SON (8) | ||
OPA1679 | Quad | SOIC (14) |
TSSOP (14) | ||
QFN (16) |
Changes from Revision D (December 2021) to Revision E (December 2022)
Changes from Revision C (April 2019) to Revision D (December 2021)
Changes from Revision B (June 2018) to Revision C (April 2019)
Changes from Revision A (May 2018) to Revision B (June 2018)
Changes from Revision * (February 2017) to Revision A (May 2018)
PIN | TYPE | DESCRIPTION | ||
---|---|---|---|---|
NAME | NO. | |||
D (SOIC) |
DBV (SOT-23) | |||
–IN | 2 | 4 | Input | Inverting input |
+IN | 3 | 3 | Input | Noninverting input |
OUT | 6 | 1 | Output | Output |
V– | 4 | 2 | Power | Negative (lowest) power supply |
V+ | 7 | 5 | Power | Positive (highest) power supply |
PIN | TYPE | DESCRIPTION | |
---|---|---|---|
NAME | NO. | ||
–IN A | 2 | Input | Inverting input, channel A |
+IN A | 3 | Input | Noninverting input, channel A |
–IN B | 6 | Input | Inverting input, channel B |
+IN B | 5 | Input | Noninverting input, channel B |
OUT A | 1 | Output | Output, channel A |
OUT B | 7 | Output | Output, channel B |
V– | 4 | Power | Negative (lowest) power supply |
V+ | 8 | Power | Positive (highest) power supply |
Thermal Pad | Thermal pad | — | For DRG (SON-8) package. Exposed thermal die pad on underside. Connect thermal die pad to V–. Solder the thermal pad to improve heat dissipation and provide specified performance. |
PIN | TYPE | DESCRIPTION | ||
---|---|---|---|---|
NAME | NO. | |||
D (SOIC) PW (TSSOP) |
RUM (QFN) | |||
–IN A | 2 | 1 | Input | Inverting input, channel A |
+IN A | 3 | 2 | Input | Noninverting input, channel A |
–IN B | 6 | 5 | Input | Inverting input, channel B |
+IN B | 5 | 4 | Input | Noninverting input, channel B |
–IN C | 9 | 8 | Input | Inverting input, channel C |
+IN C | 10 | 9 | Input | Noninverting input, channel C |
–IN D | 13 | 12 | Input | Inverting input, channel D |
+IN D | 12 | 11 | Input | Noninverting input, channel D |
NC | — | 13 | — | No connect |
NC | — | 16 | — | No connect |
OUT A | 1 | 15 | Output | Output, channel A |
OUT B | 7 | 6 | Output | Output, channel B |
OUT C | 8 | 7 | Output | Output, channel C |
OUT D | 14 | 14 | Output | Output, channel D |
V+ | 4 | 3 | Power | Positive (highest) power supply |
V– | 11 | 10 | Power | Negative (lowest) power supply |
Thermal Pad | — | Thermal pad | — | Exposed thermal die pad on underside. Connect thermal die pad to V–. Solder the thermal pad to improve heat dissipation and provide specified performance. |
MIN | MAX | UNIT | |||
---|---|---|---|---|---|
Voltage | Supply voltage, VS = (V+) – (V–) | 40 | V | ||
Input voltage | (V–) – 0.5 | (V+) + 0.5 | V | ||
Current | Input current (all pins except power-supply pins) | –10 | 10 | mA | |
Output short-circuit current(2) | Continuous | ||||
TA | Operating temperature | –55 | 125 | °C | |
TJ | Junction temperature | 150 | °C | ||
Tstg | Storage temperature | –65 | 150 | °C |
VALUE | UNIT | |||
---|---|---|---|---|
V(ESD) | Electrostatic discharge | Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001(1) | ±2000 | V |
Charged-device model (CDM), per JEDEC specification JESD22-C101(2) | ±1000 | |||
Machine model (MM)(3) | ±200 |
MIN | NOM | MAX | UNIT | |||
---|---|---|---|---|---|---|
VS | Supply voltage | Single supply | 4.5 | 36 | V | |
Dual supply | ±2.25 | ±18 | ||||
TA | Operating temperature | –40 | 125 | °C |
THERMAL METRIC(1) | OPA1677 | UNIT | ||
---|---|---|---|---|
D (SOIC) |
DBV (SOT-23) | |||
8 PINS | 5 PINS | |||
RθJA | Junction-to-ambient thermal resistance | 132.9 | 180.5 | °C/W |
RθJC(top) | Junction-to-case (top) thermal resistance | 74.0 | 78.5 | °C/W |
RθJB | Junction-to-board thermal resistance | 76.3 | 47.3 | °C/W |
ψJT | Junction-to-top characterization parameter | 24.9 | 20.4 | °C/W |
ψJB | Junction-to-board characterization parameter | 75.6 | 47.0 | °C/W |
RθJC(bot) | Junction-to-case (bottom) thermal resistance | N/A | N/A | °C/W |
THERMAL METRIC(1) | OPA1678 | UNIT | |||
---|---|---|---|---|---|
D (SOIC) |
DGK (VSSOP) | DRG (SON) | |||
8 PINS | 8 PINS | 8 PINS | |||
RθJA | Junction-to-ambient thermal resistance | 144 | 219 | 66.9 | °C/W |
RθJC(top) | Junction-to-case (top) thermal resistance | 77 | 79 | 54.5 | °C/W |
RθJB | Junction-to-board thermal resistance | 62 | 104 | 40.4 | °C/W |
ψJT | Junction-to-top characterization parameter | 28 | 15 | 1.9 | °C/W |
ψJB | Junction-to-board characterization parameter | 61 | 102 | 40.4 | °C/W |
RθJC(bot) | Junction-to-case (bottom) thermal resistance | N/A | N/A | 10.8 | °C/W |
THERMAL METRIC(1) | OPA1679 | UNIT | |||
---|---|---|---|---|---|
D (SOIC) |
PW (TSSOP) | RUM (QFN) | |||
14 PINS | 14 PINS | 16 PINS | |||
RθJA | Junction-to-ambient thermal resistance | 90 | 127 | 38.5 | °C/W |
RθJC(top) | Junction-to-case (top) thermal resistance | 55 | 47 | 34.4 | °C/W |
RθJB | Junction-to-board thermal resistance | 44 | 59 | 17.4 | °C/W |
ψJT | Junction-to-top characterization parameter | 20 | 55 | 0.6 | °C/W |
ψJB | Junction-to-board characterization parameter | 44 | 58 | 17.4 | °C/W |
RθJC(bot) | Junction-to-case (bottom) thermal resistance | N/A | N/A | 7.1 | °C/W |
PARAMETER | TEST CONDITIONS | MIN | TYP | MAX | UNIT | ||
---|---|---|---|---|---|---|---|
AUDIO PERFORMANCE | |||||||
THD+N | Total harmonic distortion + noise | G = 1, RL = 600 Ω, f = 1 kHz, VO = 3 VRMS | 0.0001% | ||||
–120 | dB | ||||||
IMD | Intermodulation distortion | G = 1 VO = 3 VRMS |
SMPTE/DIN two-tone, 4:1 (60 Hz and 7 kHz) |
0.0001% | |||
–120 | dB | ||||||
DIM 30 (3-kHz square wave and 15-kHz sine wave) |
0.0001% | ||||||
–120 | dB | ||||||
CCIF twin-tone (19 kHz and 20 kHz) |
0.0001% | ||||||
–120 | dB | ||||||
FREQUENCY RESPONSE | |||||||
GBW | Gain-bandwidth product | G = 1 | 16 | MHz | |||
SR | Slew rate | G = –1 | 9 | V/µs | |||
Full power bandwidth(1) | VO = 1 VP | 1.4 | MHz | ||||
Overload recovery time | G = –10 | 1 | µs | ||||
Channel separation (dual and quad) | f = 1 kHz | –130 | dB | ||||
NOISE | |||||||
en | Input voltage noise | f = 20 Hz to 20 kHz | 5.4 | µVPP | |||
f = 0.1 Hz to 10 Hz | 1.74 | ||||||
Input voltage noise density | f = 1 kHz | 4.5 | nV/√Hz | ||||
in | Input current noise density | f = 1 kHz | 3 | fA/√Hz | |||
OFFSET VOLTAGE | |||||||
VOS | Input offset voltage | VS = ±2.25 V to ±18 V | ±0.5 | ±2 | mV | ||
VS = ±2.25 V to ±18 V, TA = –40°C to 125°C(2) | 2 | µV/°C | |||||
PSRR | Power-supply rejection ratio | VS = ±2.25 V to ±18 V | 3 | 8 | µV/V | ||
INPUT BIAS CURRENT | |||||||
IB | Input bias current | VCM = 0 V | ±10 | pA | |||
IOS | Input offset current | VCM = 0 V | ±10 | pA | |||
INPUT VOLTAGE RANGE | |||||||
VCM | Common-mode voltage range | (V–)+0.5 | (V+) – 2 | V | |||
CMRR | Common-mode rejection ratio | 100 | 110 | dB | |||
INPUT IMPEDANCE | |||||||
Differential | 100 || 6 | MΩ || pF | |||||
Common-mode | 6000 || 2 | GΩ || pF | |||||
OPEN-LOOP GAIN | |||||||
AOL | Open-loop voltage gain | (V–) + 0.8 V ≤ VO ≤ (V+) – 0.8 V | 106 | 114 | dB | ||
OUTPUT | |||||||
VO | Output voltage | (V–) + 0.8 | (V+) – 0.8 | V | |||
IOUT | Output Current | See Section 6.8 | mA | ||||
ZO | Open-loop output impedance | f = 1 MHz | See Section 6.8 | Ω | |||
ISC | Short-circuit current(3) | ±50 | mA | ||||
CL | Capacitive load drive | 100 | pF | ||||
POWER SUPPLY | |||||||
IQ | Quiescent current (per channel) | IO = 0 A | 2 | 2.5 | mA | ||
IO = 0 A, TA = –40°C to 125°C(2) | 2.8 |
at TA = 25°C, VS = ±15 V, and RL = 2 kΩ, (unless otherwise noted)
CL = 10 pF |
VOUT = 3 VRMS | RL = 2 kΩ | Bandwidth = 80 kHz |
f = 1 kHz | RL = 2 kΩ | Bandwidth = 80 kHz |
VOUT = 3 VRMS | Gain = 1 V/V |
Gain = 1 V/V | CL = 100 pF |
Gain = +1 V/V | RF = 2 kΩ | CL = 100 pF |
G = 1 |
Gain = –10 V/V |
CL = 10 pF |
VOUT = 3 VRMS | RL = 600 Ω | Bandwidth = 80 kHz |
f = 1 kHz | RL = 600 Ω | Bandwidth = 80 kHz |
Gain = –1 V/V | CL = 100 pF |
Gain = –1 V/V | CL = 100 pF |
G = 1 |
Gain = –10 V/V |
Gain = 1 V/V |
The OPA167x devices are unity-gain stable, dual-channel and quad-channel op amps with low noise and distortion. Section 7.2 shows a simplified schematic of the OPA167x (one channel shown). These devices consist of a low-noise input stage with a folded cascode and a rail-to-rail output stage. This topology exhibits excellent noise and distortion performance across a wide range of supply voltages that are not delivered by legacy, commodity, audio operational amplifiers.
The OPA167x family has internal phase-reversal protection. Many op amps exhibit phase reversal when the input is driven beyond the linear common-mode range. This condition is most often encountered in noninverting circuits when the input is driven beyond the specified common-mode voltage range, causing the output to reverse into the opposite rail. The input of the OPA167x prevents phase reversal with excessive common-mode voltage. Instead, the appropriate rail limits the output voltage. This performance is shown in Figure 7-1.
Designers often ask questions about the capability of an operational amplifier to withstand electrical overstress. These questions tend to focus on the device inputs, but can involve the supply voltage pins or even the output pin. Each of these different pin functions have electrical stress limits determined by the voltage breakdown characteristics of the particular semiconductor fabrication process and specific circuits connected to the pin. Additionally, internal electrostatic discharge (ESD) protection is built into these circuits to protect them from accidental ESD events both before and during product assembly.
A good understanding of this basic ESD circuitry and the relevance to an electrical overstress event is helpful. Figure 7-2 illustrates the ESD circuits contained in the OPA167x (indicated by the dashed line area). The ESD protection circuitry involves several current-steering diodes connected from the input and output pins and routed back to the internal power-supply lines, where the diodes meet at an absorption device internal to the operational amplifier. This protection circuitry is intended to remain inactive during normal circuit operation.
An ESD event produces a short-duration, high-voltage pulse that is transformed into a short-duration, high-current pulse when discharging through a semiconductor device. The ESD protection circuits are designed to provide a current path around the operational amplifier core to prevent damage. The energy absorbed by the protection circuitry is then dissipated as heat.
When an ESD voltage develops across two or more amplifier device pins, current flows through one or more steering diodes. Depending on the path that the current takes, the absorption device can activate. The absorption device has a trigger, or threshold voltage, that is greater than the normal operating voltage of the OPA167x but less than the device breakdown voltage level. When this threshold is exceeded, the absorption device quickly activates and clamps the voltage across the supply rails to a safe level.
When the operational amplifier connects into a circuit (see Figure 7-2), the ESD protection components are intended to remain inactive and do not become involved in the application circuit operation. However, circumstances can arise where an applied voltage exceeds the operating voltage range of a given pin. If this condition occurs, there is a risk that some internal ESD protection circuits can turn on and conduct current. Any such current flow occurs through steering-diode paths and rarely involves the absorption device.
Figure 7-2 shows a specific example where the input voltage (VIN) exceeds the positive supply voltage (V+) by 500 mV or more. Much of what happens in the circuit depends on the supply characteristics. If V+ can sink the current, one of the upper input steering diodes conducts and directs current to V+. Excessively high current levels can flow with increasingly higher VIN. As a result, the data sheet specifications recommend that applications limit the input current to 10 mA.
If the supply is not capable of sinking the current, VIN can begin sourcing current to the operational amplifier and then take over as the source of positive supply voltage. The danger in this case is that the voltage can rise to levels that exceed the operational amplifier absolute maximum ratings.
Another common question involves what happens to the amplifier if an input signal is applied to the input when the power supplies (V+ or V–) are at 0 V. Again, this question depends on the supply characteristic when at 0 V, or at a level less than the input signal amplitude. If the supplies appear as high impedance, then the input source supplies the operational amplifier current through the current-steering diodes. This state is not a normal bias condition; most likely, the amplifier does not operate normally. If the supplies are low impedance, then the current through the steering diodes can become quite high. The current level depends on the ability of the input source to deliver current, and any resistance in the input path.
If there is any uncertainty about the ability of the supply to absorb this current, add external Zener diodes to the supply pins; see Figure 7-2. Select the Zener voltage so that the diode does not turn on during normal operation. However, the Zener voltage must be low enough so that the Zener diode conducts if the supply pin begins to rise above the safe-operating, supply-voltage level.
The electromagnetic interference (EMI) rejection ratio, or EMIRR, describes the EMI immunity of operational amplifiers. An adverse effect that is common to many operational amplifiers is a change in the offset voltage as a result of RF signal rectification. An operational amplifier that is more efficient at rejecting this change in offset as a result of EMI has a higher EMIRR and is quantified by a decibel value. Measuring EMIRR can be performed in many ways, but this document provides the EMIRR IN+, which specifically describes the EMIRR performance when the RF signal is applied to the noninverting input pin of the operational amplifier. In general, only the noninverting input is tested for EMIRR for the following three reasons:
A more formal discussion of the EMIRR IN+ definition and test method is shown in the EMI Rejection Ratio of Operational Amplifiers application report, available for download at www.ti.com.
The EMIRR IN+ of the OPA167x is plotted versus frequency in Figure 7-3. The dual and quad operational amplifier device versions have approximately identical EMIRR IN+ performance. The OPA167x unity-gain bandwidth is 16 MHz. EMIRR performance below this frequency denotes interfering signals that fall within the operational amplifier bandwidth.
Table 7-1 lists the EMIRR IN+ values for the OPA167x at particular frequencies commonly encountered in real-world applications. Applications listed in Table 7-1 can be centered on or operated near the particular frequency shown. This information can be of special interest to designers working with these types of applications, or working in other fields likely to encounter RF interference from broad sources, such as the industrial, scientific, and medical (ISM) radio band.
FREQUENCY | APPLICATION OR ALLOCATION | EMIRR IN+ |
---|---|---|
400 MHz | Mobile radio, mobile satellite, space operation, weather, radar, UHF | 36 dB |
900 MHz | GSM, radio communication and navigation, GPS (to 1.6 GHz), ISM, aeronautical mobile, UHF | 42 dB |
1.8 GHz | GSM, mobile personal comm. broadband, satellite, L-band | 52 dB |
2.4 GHz | 802.11b/g/n, Bluetooth™, mobile personal comm., ISM, amateur radio and satellite, S-band | 64 dB |
3.6 GHz | Radiolocation, aero comm./nav., satellite, mobile, S-band | 67 dB |
5 GHz | 802.11a/n, aero communication and navigation, mobile communication, space and satellite operation, C-band | 77 dB |
Figure 7-4 shows the circuit configuration for testing the EMIRR IN+. An RF source is connected to the operational amplifier noninverting input pin using a transmission line. The operational amplifier is configured in a unity-gain buffer topology with the output connected to a low-pass filter (LPF) and a digital multimeter (DMM). A large impedance mismatch at the operational amplifier input causes a voltage reflection; however, this effect is characterized and accounted for when determining the EMIRR IN+. The resulting dc offset voltage is sampled and measured by the multimeter. The LPF isolates the multimeter from residual RF signals that can interfere with multimeter accuracy. See the EMI Rejection Ratio of Operational Amplifiers application report for more details.
The OPA167x series op amps operate from ±2.25 V to ±18 V supplies while maintaining excellent performance. The OPA167x series can operate with as little as 4.5 V between the supplies and with up to 36 V between the supplies. However, some applications do not require equal positive and negative output voltage swing. With the OPA167x series, power-supply voltages are not required to be equal. For example, the positive supply can be set to 25 V with the negative supply at –5 V.
In all cases, the common-mode voltage must be maintained within the specified range. In addition, key parameters are specified over the temperature range of TA = –40°C to +85°C. Parameters that vary significantly with operating voltage or temperature are shown in Section 6.8.
Information in the following applications sections is not part of the TI component specification, and TI does not warrant its accuracy or completeness. TI’s customers are responsible for determining suitability of components for their purposes, as well as validating and testing their design implementation to confirm system functionality.
The dynamic characteristics of the OPA167x series are optimized for commonly encountered gains, loads, and operating conditions. The combination of low closed-loop gain and high capacitive loads decreases the phase margin of the amplifier, and can lead to gain peaking or oscillations. As a result, heavier capacitive loads must be isolated from the output. The simplest way to achieve this isolation is to add a small resistor (RS equal to 50 Ω, for example) in series with the output.
This small series resistor also prevents excess power dissipation if the output of the device short-circuits. For more details about analysis techniques and application circuits, see the Feedback Plots Define Op Amp AC Performance application report, available for download from the TI website (www.ti.com).
Contact microphones are useful for amplifying the sound of musical instruments that do not contain electric pickups, such as acoustic guitars and violins. Most contact microphones use a piezo element to convert vibrations in the body of the musical instrument to a voltage which can be amplified or recorded. The low noise and low input bias current of the OPA1678 make the device an excellent choice for high impedance preamplifiers for piezo elements. This preamplifier circuit provides high input impedance for the piezo element but has low output impedance for driving long cable runs. The circuit is also designed to be powered from 48-V phantom power which is commonly available in professional microphone preamplifiers and recording consoles.
A TINA-TI™ simulation schematic of the circuit below is available in the Tools and Software section of the OPA1678 or OPA1679 product folder.
In professional audio systems, phantom power is applied to the two signal lines that carry a differential audio signal from the microphone. Figure 8-2 is a diagram of the system showing 48-V phantom power applied to the differential signal lines between the piezo preamplifier output and the input of a professional microphone preamplifier.
A voltage divider is used to extract the common-mode phantom power from the differential audio signal in this type of system. The voltage at center point of the voltage divider formed by R1 and R2 does not change when audio signals are present on the signal lines (assuming R1 and R2 are matched). A Zener diode forces the voltage at the center point of R1 and R2 to a regulated voltage. The values of R1 and R2 are determined by the allowable voltage drop across these resistors from the current delivered to both op amp channels and the Zener diode. There are two power supply current pathways in parallel, each sharing half the total current of the op amp and Zener diode. Resistors R1 and R2 can be calculated using Equation 1:
A 24-V Zener diode is selected for this design, and 1 mA of current flows through the diode at idle conditions to maintain the reverse-biased condition of the Zener diode. The maximum idle power supply current of both op amp channels is 5 mA. Inserting these values into Equation 1 gives the values for R1 and R2 shown in Equation 2.
Using a value of 1.2 kΩ for resistors R1 and R2 establishes a 1-mA current through the Zener diode and properly regulate the node to 24 V. Capacitor C1 forms a low-pass filter with resistors R1 and R2 to filter the Zener diode noise and any residual differential audio signals. Mismatch in the values of R1 and R2 causes a portion of the audio signal to appear at the voltage divider center point. The corner frequency of the low-pass filter must be set below the audio band, as shown in Equation 3.
A 22-μF capacitor is selected because the capacitor meets the requirements for power supply filtering and is a widely available denomination. A 0.1-µF capacitor (C2) is added in parallel with C1 as a high-frequency bypass capacitor.
Resistors R3 and R4 provide a pathway for the input bias current of the OPA1678 while maintaining the high input impedance of the circuit. The contact microphone capacitance and the required
low-frequency response determine the values of R3 and R4. The –3-dB frequency formed by the microphone capacitance and amplifier input impedance is shown in Equation 4:
A piezo element with 8 nF of capacitance was selected for this design because the 9-kHz resonance is towards the upper end of the audible bandwidth, and is less likely to affect the frequency response of many musical instruments. The minimum value for resistors R3 and R4 is then calculated with Equation 5:
1-MΩ resistors are selected for R3 and R4 to make sure the circuit meets the design requirements for –3-dB bandwidth. The center point of resistors R3 and R4 is biased to half the supply voltage through the voltage divider formed by R5 and R6. This sets the input common-mode voltage of the circuit to a value within the input voltage range of the OPA1678. Piezo elements can produce very large voltages if the elements are struck with sufficient force. To prevent damage, the input of the OPA1678 is protected by a transient voltage suppressor (TVS) diode placed across the preamplifier inputs. The TPD1E1B04 TVS was selected due to low capacitance and the 6.4-V clamping voltage does not clamp the desired low amplitude vibration signals. Resistors R14 and R15 limit current flow into the amplifier inputs in the event that the internal protection diodes of the amplifier are forward-biased.
R7, R8, and R9 determines the gain of the preamplifier circuit. The gain of the circuit is shown in Equation 6:
Resistors R7 and R9 are selected with a value of 2 kΩ to avoid loading the output of the OPA1678 and producing distortion. The value of R8 is then calculated in Equation 7:
Capacitors C3 and C4 limit the bandwidth of the circuit so that signals outside the audio bandwidth are not amplified. The corner frequency produced by capacitors C3 and C4 is shown in Equation 8. This corner frequency must be above the desired –3-dB bandwidth point to avoid attenuating high-frequency audio signals.
C3 and C4 are 390-pF capacitors, which places the corner frequency approximately 1 decade above the desired –3-dB bandwidth point. Capacitors C3 and C4 must be NP0 or C0G type ceramic capacitors or film capacitors. Other ceramic dielectrics, such as X7R, are not suitable for these capacitors and produce distortion.
The audio signal is ac-coupled onto the microphone signal lines through capacitors C5 and C6. The value of capacitors C5 and C6 are determined by the low-frequency design requirements and the input impedance of the microphone preamplifier that connect to the output of the circuit. Equation 9 shows an approximation of the capacitor value requirements, and neglects the effects of R10, R11, R12, and R13 on the frequency response. The microphone preamplifier input impedance (RIN_MIC) uses a typical value of 4.4 kΩ for the calculation.
For simplicity, the same 22-μF capacitors selected for the power supply filtering are selected for C5 and C6 to satisfy Equation 9. At least 50-V rated capacitors must be used for C5 and C6. If polarized capacitors are used, the positive terminal must be oriented towards the microphone preamplifier. Resistors R10 and R11 isolate the op amp outputs from the capacitance of long cables that can cause instability. R12 and R13 discharge ac-coupling capacitors C4 and C5 when phantom power is removed.
The frequency response of the preamplifier circuit is shown in Figure 8-3. The –3-dB frequencies are 15.87 Hz and 181.1 kHz, which meet the design requirements. The gain within the passband of the circuit is 18.9 dB, slightly less than the design goal of 20 dB. The reduction in gain is a result of the voltage division between the output resistors of the piezo preamplifier circuit and the input impedance of the microphone preamplifier. The A-weighted noise of the circuit (referred to the input) is 842.2 nVRMS or –119.27 dBu.
The noise and distortion performance of the OPA167x family of amplifiers is exceptional in applications with high source impedances, which makes these devices a viable choice in preamplifier circuits for moving magnet (MM) phono cartridges. Figure 8-4 shows a preamplifier circuit for MM cartridges with 40 dB of gain at 1 kHz.
The preamplifier circuit shown in Figure 8-5 operates the OPA1678 as a transimpedance amplifier that converts the output current from the electret microphone internal JFET into a voltage. Resistor R4 determines the gain of the circuit. Resistors R2 and R3 bias the input voltage to half the power supply voltage for proper functionality on a single-supply.
Figure 8-6 shows the BUF634A buffer inside the feedback loop of the OPA1678 to increase the available output current for low-impedance headphones. If the BUF634A is used in wide-bandwidth mode, no additional components besides the feedback resistors are required to maintain loop stability.
Figure 8-7 shows the OPA1678 used as an integrator that drives the reference pin of the INA1650, which forces the output dc voltage to 0 V. This configuration is an alternative to large ac-coupling capacitors that can distort at high output levels. The low input bias current and low input offset voltage of the OPA1678 make the device an excellent choice for integrator applications.
The OPA167x devices are specified for operation from 4.5 V to 36 V (±2.25 V to ±18 V); many specifications apply from –40°C to +85°C. Parameters that can exhibit significant variance with regard to operating voltage or temperature are shown in Section 6.8. Applications with noisy or high-impedance power supplies require decoupling capacitors close to the device pins. In most cases, 0.1-µF capacitors are adequate.
For best operational performance of the device, use good printed-circuit board (PCB) layout practices, including:
The OPA167x series op amps are capable of driving 2-kΩ loads with a power-supply voltage up to ±18 V and full operating temperature range. Internal power dissipation increases when operating at high supply voltages. Copper leadframe construction used in the OPA167x series op amps improves heat dissipation compared to conventional materials. Circuit board layout can also help minimize junction temperature rise. Wide copper traces help dissipate the heat by acting as an additional heat sink. Temperature rise can be further minimized by soldering the devices to the circuit board rather than using a socket.
PSpice® for TI is a design and simulation environment that helps evaluate performance of analog circuits. Create subsystem designs and prototype solutions before committing to layout and fabrication, reducing development cost and time to market.
TINA-TI™ simulation software is a simple, powerful, and easy-to-use circuit simulation program based on a SPICE engine. TINA-TI simulation software is a free, fully-functional version of the TINA™ software, preloaded with a library of macromodels, in addition to a range of both passive and active models. TINA-TI simulation software provides all the conventional dc, transient, and frequency domain analysis of SPICE, as well as additional design capabilities.
Available as a free download from the Design tools and simulation web page, TINA-TI simulation software offers extensive post-processing capability that allows users to format results in a variety of ways. Virtual instruments offer the ability to select input waveforms and probe circuit nodes, voltages, and waveforms, creating a dynamic quick-start tool.
These files require that either the TINA software or TINA-TI software be installed. Download the free TINA-TI simulation software from the TINA-TI™ software folder.
Speed up your op amp prototyping and testing with the DIP-Adapter-EVM, which provides a fast, easy and inexpensive way to interface with small, surface-mount devices. Connect any supported op amp using the included Samtec terminal strips or wire them directly to existing circuits. The DIP-Adapter-EVM kit supports the following industry-standard packages: D or U (SOIC-8), PW (TSSOP-8), DGK (VSSOP-8), DBV (SOT-23-6, SOT-23-5 and SOT-23-3), DCK (SC70-6 and SC70-5), and DRL (SOT563-6).
The DIYAMP-EVM is a unique evaluation module (EVM) that provides real-world amplifier circuits, enabling the user to quickly evaluate design concepts and verify simulations. This EVM is available in three industry-standard packages (SC70, SOT23, and SOIC) and 12 popular amplifier configurations, including amplifiers, filters, stability compensation, and comparator configurations for both single and dual supplies.
TI reference designs are analog solutions created by TI’s precision analog applications experts. TI reference designs offer the theory of operation, component selection, simulation, complete PCB schematic and layout, bill of materials, and measured performance of many useful circuits. TI reference designs are available online at https://www.ti.com/reference-designs.
The filter design tool is a simple, powerful, and easy-to-use active filter design program. The filter design tool allows the user to create optimized filter designs using a selection of TI operational amplifiers and passive components from TI's vendor partners.
Available as a web-based tool from the Design tools and simulation web page, the filter design tool allows the user to design, optimize, and simulate complete multistage active filter solutions within minutes.
The following documents are relevant to using the OPA167x, and are recommended for reference. All are available for download at www.ti.com unless otherwise noted.
To receive notification of documentation updates, navigate to the device product folder on ti.com. Click on Subscribe to updates to register and receive a weekly digest of any product information that has changed. For change details, review the revision history included in any revised document.
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TINA-TI™ and TI E2E™ are trademarks of Texas Instruments.
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This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. |
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. |
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