7.3.1 MOSFET Gate Drive

The output-driver sinking current is approximately 200 mA and is designed to drive P-channel power FETs. When the driver pulls the gate charge of the FET, it is controlling to 8  V (typical), the drive current folds back to a low level so that high-power dissipation only occurs during the turn-on period of the FET. This feature is particularly valuable when turning on a FET at high input voltages, where leaving the gate drive current on would otherwise cause unacceptable power dissipation.

7.3.2 Undervoltage Lockout Protection

Undervoltage lockout (UVLO) protection ensures proper start-up of the device only when the input voltage has exceeded minimum operating voltage. UVLO protection incorporates hysteresis that eliminates hiccup starting in cases where input supply impedance is high.

Figure 23. Undervoltage Lockout

Undervoltage protection ensures proper startup of the device only when the input voltage has exceeded minimum operating voltage. The UVLO level is measured at the V_DD pin with respect to GND. Start-up voltage is typically 4.3 V with approximately 200 mV of hysteresis. The device shuts off at a nominal 4.1 V. As shown in Figure 23, when the input V_DD voltage rises to 4.3 V, the 1.3-V comparator threshold voltage is exceeded a d a RUN signal occurs. Feedback from the output closes the switch and shunts the 200-kΩ resistor, so that an approximately 200-mV lower voltage, or 4.1 V, is required before the device shuts down.

7.3.3 Selecting the Operating Frequency

The operating frequency of the controller is determined by an external resistor R_RC that is connected from the RC pin to VDD and a capacitor attached from the RC pin to ground. This connection and the two oscillator comparators inside the device are shown in Figure 24. The oscillator frequency can be calculated using Equation 1.

Equation 1.

where

• f_SW = clock frequency
• R_RC = timing resistor value in Ω
• C_RC = timing capacitor value in F

R_RC must be kept large enough that the current through it does not exceed 750 μA when the internal switch (shown in Figure 24) is discharging the timing capacitor. This condition may be expressed using Equation 2.

Equation 2.

Figure 24. Oscillator Functional Diagram

7.3.4 Synchronizing the Oscillator

Figure 24 shows the functional diagram of the TPS40200-Q1 oscillator. When synchronizing the oscillator to an external clock, the RC pin must be pulled below 150 mV for 20 ns or more. The external clock frequency must also be higher than the free-running frequency of the converter. When synchronizing the controller, if the RC pin is held low for an excessive amount of time, erratic operation may occur. The maximum amount of time that the RC pin should be held low is the lesser of the following:

• 50% of a nominal output pulse, or
• 10% of the period of the synchronization frequency

Under circumstances where the input voltage is high and the duty cycle is less than 50%, a Schottky diode connected from the RC pin to an external clock may be used to synchronize the oscillator (see Figure 25). The cathode of the diode is connected to the RC pin. The trip point of the oscillator is set by an internal voltage divider to be 1/10 of the input voltage. The clock signal must have an amplitude higher than this trip point. When the clock goes low, it allows the reset current to restart the RC ramp , synchronizing the oscillator to the external clock. This provides a simple, single-component method for clock synchronization.

Figure 25. Diode-Connected Synchronization

7.3.5 Current-Limit Resistor Selection

As shown in Figure 28, a resistor in series with the power MOSFET sets the overcurrent protection level. Use a low-inductance resistor to avoid ringing signals and nuisance tripping. When the FET is on and the controller senses 100 mV or more drop from the V_DD pin to the ISNS pin, an overcurrent condition is declared. When this happens, the FET is turned off and, as shown in Figure 26, the soft-start capacitor is discharged. When the soft-start capacitor reaches a level below 150 mV, the converter clears the overcurrent condition flag and attempts to restart. If the condition that caused the overcurrent event to occur is still present on the output of the converter (see Figure 27), another overcurrent condition is declared and the process repeats indefinitely. Figure 27 shows the soft-start capacitor voltage during an extended output fault condition. The overall duty cycle of current conduction during a persistent fault is approximately 2%.

Figure 26. Current-Limit Reset

Figure 27. Typical Soft-Start Capacitor and V_OUT During Overcurrent

If necessary, a small RC filter can be added to the current sensing network to reduce nuisance tripping due to noise pickup. This filter can also be used to trim the overcurrent trip point to a higher level with the addition of a single resistor (see Figure 28). The nominal overcurrent trip point using the circuit of Figure 28 is described using Equation 3.

Equation 3.

where

• I_OC = overcurrent trip point, peak current in the inductor
• V_ILIM = overcurrent threshold voltage for the TPS40200-Q1, typically 100 mV
• R_ILIM = value of the current sense resistor in Ω
• R_F1 and R_F2 = values of the scaling resistors in Ω

The value of the capacitor is determined by the nominal pulse width of the converter and the values of the scaling resistors R_F1 and R_F2. It is best not to have the time constant of the filter longer than the nominal pulse width of the converter, otherwise a substantial increase in the overcurrent trip point occurs. Using this constraint, the capacitor value may be bounded using Equation 4.

Equation 4.

where

• C_f = value of the current limit filter capacitor in F
• V_O = output voltage of the converter
• V_IN = input voltage to the converter
• f_SW = converter switching frequency
• R_F1 and R_F2 = values of the scaling resistors in Ω

Figure 28. Current-Limit Adjustment

7.3.6 Calculating the Soft-Start Time

An external capacitor (C_SS) connected from the SS pin to ground controls the TPS40200-Q1 soft-start interval. An internal charging resistor connected to V_DD produces a rising reference voltage that is connected through a 700-mV offset to the reference input of the TPS40200-Q1 error amplifier. When the soft-start capacitor voltage (V_CSS) is below 150 mV, there is no switching activity. When V_CSS rises above the 700-mV offset, the error amplifier starts to follow V_SST – 700 mV and uses this rising voltage as a reference. When V_CSS reaches 1.4 V, the internal reference takes over, and further increases have no effect. An advantage of initiating a slow start in this fashion is that the controller cannot overshoot, because its output follows a scaled version of the controller reference voltage. A conceptual drawing of the circuit that produces these results is shown in Figure 29. A consequence of the 700-mV offset is that the controller does not start switching until the V_CSS has charged up to 700 mV. The output remains at 0 V during the resulting delay. When V_CCS exceeds the 700-mV offset, the TPS40200-Q1 output follows the soft-start time constant. Once above 1.4 V, the 700-mV internal reference takes over, and normal operation begins.

Figure 29. Soft-Start Circuit

The slow-start time should be longer (slower) than the time constant of the output LC filter. This time constraint may be expressed as:

Equation 5.

The calculation of the soft-start interval is simply the time it takes the RC network to exponentially charge from 0 V to 1.4 V. An internal 105-kΩ charging resistor is connected from the SS pin to V_SST. For applications where the voltage is above 8 V, an internal regulator clamps the maximum charging voltage to 8 V.

The result of this is a formula for the start up time, as given by Equation 6.

Equation 6.

where

• t_SS = required soft-start time in seconds
• C_SS = soft-start capacitor value in F
• R_c = internal soft-start charging resistor (105 kΩ nominal)
• V_SST = input voltage up to a maximum of 8 V

7.3.7 Voltage Setting and Modulator Gain

Since the input current to the error amplifier is negligible, the feedback impedance can be selected over a wide range. Knowing that the reference voltage is 708 mV, choose a convenient value for R1 and then calculate the value of R2 using Equation 7.

Equation 7.

Figure 30. System Gain Elements

The error amplifier has a DC open-loop gain of at least 60 dB with a minimum of a 1.5-MHz gain bandwidth product, which gives the user flexibility with respect to the type of feedback compensation to use for the particular application. The gain selected by the user at the crossover frequency is set to provide an overall unity gain for the system. The crossover frequency should be selected so that the error amplifier open-loop gain is high with respect to the required closed-loop gain. This ensures that the amplifier response is determined by the passive feedback elements.

7.4.1 Operation Near Minimum Input Voltage

The TPS40200-Q1 is designed to operate with input voltages above 4.5 V. The typical V_DD UVLO threshold is 4.25 V and the device may operate at input voltages down to the UVLO voltage. At input voltages below the actual UVLO voltage, the device will not switch. When V_VDD passes the UVLO threshold, the device will become active. Switching is enabled and the soft-start sequence is initiated. The TPS40200-Q1 will ramp up the output voltage at the rate determined by the external capacitor at the soft-start pin.

7.4.2 Operation With SS Pin

The SS pin has a 150-mV threshold, which can be used to disable the TPS40200-Q1. With SS forced below this threshold voltage, the device is disabled and switching is inhibited even if V_VDD is above its UVLO threshold. If the SS voltage is allowed to increase above the threshold while V_VDD is above its UVLO threshold, the device becomes active. Switching is enabled and the soft-start sequence is initiated. The TPS40200-Q1 will ramp up the output voltage at the rate determined by the external capacitor at the soft-start pin.