Refer to the PDF data sheet for device specific package drawings
The ADS1235 is a precision, 7200-SPS, delta-sigma (ΔΣ) analog-to-digital converter (ADC) with an integrated programmable gain amplifier (PGA). This device also includes diagnostic features such as PGA overrange and reference monitors. The ADC provides high-accuracy, zero-drift conversion data for high-precision equipment, including weigh scales, strain gauges, and resistive pressure sensors.
The ADC has signal and reference multiplexers that support three differential signal inputs and two reference inputs. The ADC also includes a low-noise PGA that provides gains of 1, 64, and 128. The ADC also has a 24-bit ΔΣ modulator and programmable digital filter.
The high-impedance inputs (1 GΩ) of the PGA reduce measurement error that is caused by sensor loading.
The ADC supports ac-bridge excitation to remove the drift errors from the sensor wiring. The ADC provides the clock control signals for the ac-excitation operation.
The flexible digital filter is programmable for single-cycle settled conversions, and provides 50-Hz and 60-Hz line cycle rejection at the same time.
The ADS1235 is available in a 5-mm × 5-mm QFN package, and is specified across the –40°C to +125°C temperature range.
PART NUMBER | PACKAGE | BODY SIZE (NOM) |
---|---|---|
ADS1235 | VQFN (32) | 5.0 mm × 5.0 mm |
DATE | REVISION | NOTES |
---|---|---|
October 2018 | * | Initial release. |
PIN | TYPE | DESCRIPTION | |
---|---|---|---|
NO. | NAME | ||
1 | NC | — | No connection; float or connect to AVSS |
2 | CAPP | Analog output | PGA output P; connect a 4.7-nF C0G dielectric capacitor across CAPP and CAPN |
3 | CAPN | Analog output | PGA output N; connect a 4.7-nF C0G dielectric capacitor across CAPP and CAPN |
4 | AVDD | Analog | Positive analog power supply |
5 | AVSS | Analog | Negative analog power supply |
6 | NC | — | No connection - solder the pin for mechanical support, float or connect to DGND |
7 | PWDN | Digital input | Power down, active low |
8 | RESET | Digital input | Reset, active low |
9 | START | Digital input | Start conversion control, active high |
10 | CS | Digital input | Serial interface chip select, active low |
11 | SCLK | Digital Input | Serial interface shift clock |
12 | DIN | Digital Input | Serial interface data input |
13 | DRDY | Digital output | Data ready indicator, active low |
14 | DOUT/DRDY | Digital output | Dual function serial interface data output and active-low data ready indicator |
15 | BYPASS | Analog output | Internal subregulator bypass; connect a 1-µF capacitor to DGND |
16 | DGND | Digital | Digital ground |
17 | DVDD | Digital | Digital power supply |
18 | CLKIN | Digital input | 1) Internal oscillator: connect to DGND, 2) External clock: connect clock input |
19-24 | NC | — | No connection - solder the pin for mechanical support, float or connect to DGND |
25 | AIN5 | Analog input | Analog input 5 |
26 | AIN4 | Analog input | Analog input 4 |
27 | AIN3 | Analog input/output | Analog input 3, GPIO3, ACX2 |
28 | AIN2 | Analog input/output | Analog input 2, GPIO2, ACX1 |
29 | AIN1 | Analog input/output | Analog input 1, GPIO1, ACX2, Reference input 1 negative |
30 | AIN0 | Analog input/output | Analog input 0, GPIO0, ACX1, Reference input 1 positive |
31 | REFN0 | Analog input/output | Reference input 0 negative |
32 | REFP0 | Analog input/output | Reference input 0 positive |
— | Thermal Pad | — | Exposed thermal pad - solder the pad for mechanical support; connect to AVSS. |
MIN | MAX | UNIT | ||
---|---|---|---|---|
Power supply voltage | AVDD to AVSS | –0.3 | 7 | V |
AVSS to DGND | –3 | 0.3 | ||
DVDD to DGND | –0.3 | 7 | ||
Analog input voltage | AINx, REFP0, REFN0 | AVSS – 0.3 | AVDD + 0.3 | V |
Digital input voltage | CS, SCLK, DIN, DOUT/DRDY, DRDY, START, RESET, PWDN, CLKIN | DGND – 0.3 | DVDD + 0.3 | V |
Input Current | Continuous, all pins except power-supply pins(2) | –10 | 10 | mA |
Temperature | Junction, TJ | 150 | °C | |
Storage, Tstg | –60 | 150 | °C |
VALUE | UNIT | |||
---|---|---|---|---|
V(ESD) | Electrostatic discharge | Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001(1) | ±2000 | V |
Charged-device model (CDM), per JEDEC specification JESD22-C101(2) | ±500 |
MIN | NOM | MAX | UNIT | |||
---|---|---|---|---|---|---|
POWER SUPPLY | ||||||
Analog power supply | AVDD to AVSS | 4.75 | 5 | 5.25 | V | |
AVSS to DGND | –2.6 | 0 | ||||
Digital power supply | DVDD to DGND | 2.7 | 5.25 | V | ||
ANALOG INPUTS | ||||||
V(AINx) | Absolute input voltage | PGA mode | See Equation 3 | V | ||
PGA bypassed | AVSS – 0.1 | AVDD + 0.1 | ||||
VIN | Differential input voltage | VIN = VAINP – VAINN | ±VREF / Gain | See (1) | V | |
VOLTAGE REFERENCE INPUTS | ||||||
VREF | Differential reference voltage | VREF = V(REFPx) – V(REFNx) | 0.9 | AVDD – AVSS | V | |
V(REFNx) | Negative reference voltage | AVSS – 0.05 | V(REFPx) – 0.9 | V | ||
V(REFPx) | Positive reference voltage | V(REFNx) + 0.9 | AVDD + 0.05 | V | ||
EXTERNAL CLOCK | ||||||
fCLK | Frequency | 1 | 7.3728 | 8 | MHz | |
Duty cycle | 40% | 60% | ||||
GENERAL-PURPOSE INPUTS/OUTPUTS (GPIOs) | ||||||
Input voltage | AVSS | AVDD | V | |||
DIGITAL INPUTS (other than GPIOs) | ||||||
Input voltage | DGND | DVDD | V | |||
TEMPERATURE | ||||||
TA | Operating ambient temperature | –40 | 125 | °C |
THERMAL METRIC(1) | ADS1235 | UNIT | |
---|---|---|---|
VQFN (RHB) | |||
32 PINS | |||
RθJA | Junction-to-ambient thermal resistance | 28.6 | °C/W |
RθJC(top) | Junction-to-case (top) thermal resistance | 17.0 | °C/W |
RθJB | Junction-to-board thermal resistance | 9.7 | °C/W |
ψJT | Junction-to-top characterization parameter | 0.2 | °C/W |
ψJB | Junction-to-board characterization parameter | 9.7 | °C/W |
RθJC(bot) | Junction-to-case (bottom) thermal resistance | 0.7 | °C/W |
PARAMETER | TEST CONDITIONS | MIN | TYP | MAX | UNIT | ||
---|---|---|---|---|---|---|---|
ANALOG INPUTS | |||||||
Absolute input current | PGA mode, V(AINx) = 2.5 V | 4 | 6 | nA | |||
PGA bypass | 200 | ||||||
Absolute input current drift | 0.01 | nA/°C | |||||
Differential input current | PGA mode, VIN = 39 mV | ±0.1 | nA | ||||
PGA mode, VIN = 2.5 V | –3 | ±1 | 3 | ||||
PGA and Chop modes, VIN = 2.5 V(5) | ±5 | ||||||
PGA bypass, VIN = 2.5 V | ±40 | ||||||
Differential input current drift | 0.05 | nA/°C | |||||
Differential input impedance | PGA mode | 1 | GΩ | ||||
PGA bypass | 50 | MΩ | |||||
Crosstalk | 0.1 | µV/V | |||||
PGA | |||||||
Gain settings | 1, 64, 128 | V/V | |||||
Antialias filter frequency | CCAPP, CAPN = 4.7 nF | 60 | kHz | ||||
Output voltage monitor | Low threshold | AVSS + 0.2 | V | ||||
High threshold | AVDD – 0.2 | ||||||
PERFORMANCE | |||||||
Resolution | No missing codes | 24 | Bits | ||||
Equivalent input noise density | Gain = 64 and 128 | 8 | nV/√Hz | ||||
DR | Data rate | 2.5 | 7200 | SPS | |||
Noise performance | See Table 1 | ||||||
INL | Integral non-linearity | Gain = 1, 64 and 128 | –10 | ±2 | 10 | ppmFSR | |
VOS | Offset voltage | TA = 25°C | –350 / Gain – 5 | ±50 / Gain | 350 / Gain + 5 | µV | |
Chop mode | –0.5 / Gain – 0.05 | ±0.2 / Gain | 0.5 / Gain + 0.05 | ||||
After calibration | on the level of noise | ||||||
Offset voltage drift | Gain = 1 | 100 | 350 | nV/°C | |||
Gain = 64 and 128 | 10 | 50 | |||||
Chop mode, gain = 1, 64 and 128 | 1 | 5 | |||||
GE | Gain error | TA = 25°C | –0.5% | ±0.05% | 0.5% | ||
After calibration | on the level of noise | ||||||
Gain drift | 0.5 | 4 | ppm/°C | ||||
NMRR | Normal-mode rejection ratio(1) | See Table 5 | |||||
CMRR | Common-mode rejection ratio(2) | Data rate = 20 SPS | 130 | dB | |||
Data rate = 400 SPS | 105 | 115 | |||||
PSRR | Power-supply rejection ratio(3) | AVDD and AVSS | 90 | 100 | dB | ||
DVDD | 100 | 120 | |||||
INTERNAL OSCILLATOR | |||||||
fCLK | Frequency | 7.3728 | MHz | ||||
Accuracy | –2% | ±0.5% | 2% | ||||
VOLTAGE REFERENCE INPUTS | |||||||
Reference input current | 500 | nA | |||||
Input current vs voltage | 100 | nA/V | |||||
Input current drift | 0.1 | nA//V/°C | |||||
Input impedance | Differential | 5 | MΩ | ||||
Low voltage monitor | Threshold low | 0.4 | 0.6 | V | |||
TEMPERATURE SENSOR | |||||||
Sensor voltage | TA = 25°C | 122.4 | mV | ||||
Temperature coefficient | 420 | µV/°C | |||||
GENERAL-PURPOSE INPUTS/OUTPUTS (GPIOs)(6) | |||||||
VOL | Low-level output voltage | IOL = –1 mA | 0.2 · AVDD | V | |||
VOH | High-level output voltage | IOH = 1 mA | 0.8 · AVDD | V | |||
VIL | Low-level input voltage | 0.3 · AVDD | V | ||||
VIH | High-level input voltage | 0.7 · AVDD | V | ||||
Input hysteresis | 0.5 | V | |||||
DIGITAL INPUTS/OUTPUTS (Other than GPIOs) | |||||||
VOL | Low-level output voltage | IOL = –1 mA | 0.2 · DVDD | V | |||
IOL = –8 mA | 0.2 · DVDD | ||||||
VOH | High-level output voltage | IOH = 1 mA | 0.8 · DVDD | V | |||
IOH = 8 mA | 0.75 · DVDD | ||||||
VIL | Low-level input voltage | 0.3 · DVDD | V | ||||
VIH | High-level input voltage | 0.7 · DVDD | V | ||||
Input hysteresis | 0.1 | V | |||||
Input leakage | VIH or VIL | –10 | 10 | µA | |||
POWER SUPPLY | |||||||
IAVDD
IAVSS |
Analog supply current | PGA bypass | 2.7 | 4.5 | mA | ||
PGA mode, gain = 64 and 128 | 4.3 | 6.5 | |||||
Power-down mode | 2 | 8 | µA | ||||
IDVDD | Digital supply current | 0.4 | 0.65 | mA | |||
Power-down mode(4) | 30 | 50 | µA | ||||
PD | Power dissipation | PGA mode | 23 | 35 | mW | ||
Power-down mode | 0.1 | 0.2 |
MIN | MAX | UNIT | ||
---|---|---|---|---|
SERIAL INTERFACE | ||||
td(CSSC) | Delay time, first SCLK rising edge after CS falling edge(1) | 50 | ns | |
tsu(DI) | Setup time, DIN valid before SCLK falling edge | 25 | ns | |
th(DI) | Hold time, DIN valid after SCLK falling edge | 25 | ns | |
tc(SC) | SCLK period(2) | 97 | 106 | ns |
tw(SCH), tw(SCL) | Pulse duration, SCLK high or low | 40 | ns | |
td(SCCS) | Delay time, last SCLK falling edge before CS rising edge | 50 | ns | |
tw(CSH) | Pulse duration, CS high to reset interface | 25 | ns | |
td(SCIR) | Delay time, SCLK high or low to force interface auto-reset | 65540 | 1/fCLK | |
RESET | ||||
tw(RSTL) | Pulse duration, RESET low | 4 | 1/fCLK | |
CONVERSION CONTROL | ||||
tw(STH) | Pulse duration, START high | 4 | 1/fCLK | |
tw(STL) | Pulse duration, START low | 4 | 1/fCLK | |
tsu(DRST) | Setup time, START low or STOP command after DRDY low to stop next conversion (Continuous-conversion mode) | 100 | 1/fCLK | |
th(DRSP) | Hold time, START low or STOP command after DRDY low to continue next conversion (Continuous-conversion mode) | 150 | 1/fCLK |
PARAMETER | MIN | TYP | MAX | UNIT | |
---|---|---|---|---|---|
SERIAL INTERFACE | |||||
tw(DRH) | Pulse duration, DRDY high | 16 | 1/fCLK | ||
tp(CSDO) | Propagation delay time, CS falling edge to DOUT/DRDY driven | 0 | 50 | ns | |
tp(SCDO1) | Propagation delay time, SCLK rising edge to valid DOUT/DRDY | 40 | ns | ||
th(SCDO1) | Hold time, SCLK rising edge to invalid data on DOUT/DRDY | 0 | ns | ||
th(SCDO2) | Hold time, last SCLK falling edge of operation to invalid data on DOUT/DRDY | 15 | ns | ||
tp(SCDO2) | Propagation delay time, last SCLK falling edge to valid data ready function on DOUT/DRDY | 110 | ns | ||
tp(CSDOZ) | Propagation delay time, CS rising edge to DOUT/DRDY high impedance | 50 | ns | ||
RESET | |||||
tp(RSCN) | Propagation delay time, RESET rising edge or RESET command to start of conversion | 512 | 1/fCLK | ||
tp(PRCM) | Propagation delay time, power-on threshold voltage to ADC communication | 216 | 1/fCLK | ||
tp(CMCN) | Propagation delay time, ADC communication to conversion start | 512 | 1/fCLK | ||
AC EXCITATION | |||||
td(ACX) | Delay time, phase-to-phase blanking period | 8 | 1/fCLK | ||
tc(ACX) | ACX period | 2 | tSTDR | ||
CONVERSION CONTROL | |||||
tp(STDR) | Propagation delay time, START high or START command to DRDY high | 2 | 1/fCLK |
After calibration, shorted input |
Shorted input, 30 units |
After calibration |
30 units |
20 SPS, Sinc4 mode |
7200 SPS, Gain = 128, 8192 data points, shorted inputs |
Chop Mode, Gain = 128 |
Gain = 128 |
VREFN = AVSS |
30 units |
30 units |
Chop mode, after calibration, shorted input |
Chop mode, shorted input, 30 units |
After calibration |
After calibration |
7200 SPS, Sinc1 mode |
20 SPS, Gain = 128, 256 data points, shorted inputs |
30 Units |
Gain = 128 |
PGA Bypass (gain = 1) |
Chop Mode, Gain = 128 |
30 units |
30 units |
30 units |
The ADS1235 noise performance depends on the ADC configuration: data rate, PGA gain, digital filter configuration, and chop mode. The combination of the parameters affect noise performance. Two significant factors affecting noise performance are data rate and PGA gain. Since the profile of noise is predominantly white (flat vs frequency), decreasing the data rate proportionally decreases bandwidth and therefore, decreases total noise. Since the noise of the PGA is lower than that of the modulator, increasing the gain deceases overall conversion noise when treated as an input-referred quantity. Noise performance also depends on the digital filter and chop mode. As the order of the digital filter increases, the noise bandwidth correspondingly decreases resulting in decreased noise. Further, as a result of two-point data averaging performed in chop mode, noise decreases by √2 compared to normal operation.
Table 1 shows noise performance in units of μVRMS (RMS = root mean square) and in units of effective resolution (bits) under the conditions listed. The values in parenthesis are peak-to-peak values (µV) and noise free resolution (bits). Noise-free resolution is the resolution of the ADC with no code flicker. The noise-free resolution data are calculated based on the peak-to-peak noise data.
The effective resolution data listed in the tables are calculated using Equation 1:
where
The data shown in the noise performance table represent typical ADC performance at TA = 25°C. The noise-performance data are the standard deviation and peak-to-peak computations of the ADC data. The noise data are acquired with inputs shorted, based on consecutive ADC readings for a period of ten seconds or 8192 data points, whichever occurs first. Because of the statistical nature of noise, repeated noise measurements may yield higher or lower noise performance results.
Noise (µVRMS, (µVPP)) | Effective Resolution (bits), (Noise-Free Resolution (bits)) | ||||||
---|---|---|---|---|---|---|---|
DATA RATE | FILTER | GAIN = 1 | GAIN = 64 | GAIN = 128 | GAIN = 1 | GAIN = 64 | GAIN = 128 |
2.5 SPS | FIR | 0.21 (0.6) | 0.008 (0.028) | 0.011 (0.042) | 24 (23.8) | 24 (22.4) | 22.7 (20.8) |
2.5 SPS | Sinc1 | 0.12 (0.3) | 0.009 (0.037) | 0.008 (0.033) | 24 (24) | 24 (22) | 23.2 (21.2) |
2.5 SPS | Sinc2 | 0.15 (0.3) | 0.007 (0.023) | 0.006 (0.021) | 24 (24) | 24 (22.7) | 23.7 (21.8) |
2.5 SPS | Sinc3 | 0.15 (0.3) | 0.007 (0.023) | 0.005 (0.014) | 24 (24) | 24 (22.7) | 24 (22.4) |
2.5 SPS | Sinc4 | 0.15 (0.3) | 0.005 (0.019) | 0.006 (0.019) | 24 (24) | 24 (23) | 23.7 (22) |
5 SPS | FIR | 0.29 (0.89) | 0.013 (0.051) | 0.013 (0.051) | 24 (23.2) | 23.6 (21.5) | 22.5 (20.5) |
5 SPS | Sinc1 | 0.15 (0.3) | 0.015 (0.051) | 0.01 (0.044) | 24 (24) | 23.4 (21.5) | 22.9 (20.8) |
5 SPS | Sinc2 | 0.17 (0.6) | 0.012 (0.047) | 0.009 (0.035) | 24 (23.8) | 23.7 (21.7) | 23 (21.1) |
5 SPS | Sinc3 | 0.12 (0.6) | 0.011 (0.047) | 0.008 (0.037) | 24 (23.8) | 23.7 (21.7) | 23.1 (21) |
5 SPS | Sinc4 | 0.088 (0.3) | 0.007 (0.028) | 0.007 (0.03) | 24 (24) | 24 (22.4) | 23.3 (21.3) |
10 SPS | FIR | 0.36 (1.5) | 0.022 (0.11) | 0.02 (0.096) | 24 (22.5) | 22.8 (20.5) | 21.9 (19.6) |
10 SPS | Sinc1 | 0.28 (0.89) | 0.015 (0.065) | 0.016 (0.082) | 24 (23.2) | 23.3 (21.2) | 22.2 (19.9) |
10 SPS | Sinc2 | 0.26 (0.89) | 0.015 (0.061) | 0.013 (0.065) | 24 (23.2) | 23.3 (21.3) | 22.5 (20.2) |
10 SPS | Sinc3 | 0.26 (0.6) | 0.014 (0.065) | 0.011 (0.047) | 24 (23.8) | 23.4 (21.2) | 22.7 (20.7) |
10 SPS | Sinc4 | 0.24 (0.6) | 0.013 (0.056) | 0.01 (0.042) | 24 (23.8) | 23.6 (21.4) | 22.9 (20.8) |
16.6 SPS | Sinc1 | 0.41 (1.8) | 0.025 (0.12) | 0.022 (0.12) | 24 (22.2) | 22.6 (20.3) | 21.8 (19.4) |
16.6 SPS | Sinc2 | 0.32 (1.5) | 0.018 (0.089) | 0.018 (0.096) | 24 (22.5) | 23 (20.8) | 22 (19.6) |
16.6 SPS | Sinc3 | 0.3 (1.2) | 0.017 (0.079) | 0.018 (0.091) | 24 (22.8) | 23.1 (20.9) | 22.1 (19.7) |
16.6 SPS | Sinc4 | 0.23 (1.2) | 0.015 (0.084) | 0.014 (0.072) | 24 (22.8) | 23.3 (20.8) | 22.4 (20) |
20 SPS | FIR | 0.51 (2.1) | 0.032 (0.16) | 0.029 (0.16) | 24 (22) | 22.2 (19.9) | 21.3 (18.9) |
20 SPS | Sinc1 | 0.44 (2.1) | 0.025 (0.13) | 0.026 (0.13) | 24 (22) | 22.6 (20.2) | 21.5 (19.2) |
20 SPS | Sinc2 | 0.36 (1.2) | 0.02 (0.12) | 0.02 (0.1) | 24 (22.8) | 22.9 (20.4) | 21.9 (19.5) |
20 SPS | Sinc3 | 0.32 (1.5) | 0.017 (0.089) | 0.018 (0.096) | 24 (22.5) | 23.1 (20.8) | 22 (19.6) |
20 SPS | Sinc4 | 0.3 (1.2) | 0.017 (0.084) | 0.018 (0.1) | 24 (22.8) | 23.1 (20.8) | 22.1 (19.6) |
50 SPS | Sinc1 | 0.63 (3.6) | 0.04 (0.25) | 0.038 (0.23) | 23.7 (21.2) | 21.9 (19.2) | 21 (18.4) |
50 SPS | Sinc2 | 0.57 (3) | 0.033 (0.21) | 0.032 (0.18) | 23.9 (21.5) | 22.2 (19.5) | 21.2 (18.7) |
50 SPS | Sinc3 | 0.53 (2.4) | 0.03 (0.19) | 0.03 (0.17) | 24 (21.8) | 22.3 (19.7) | 21.3 (18.8) |
50 SPS | Sinc4 | 0.49 (2.4) | 0.028 (0.15) | 0.026 (0.16) | 24 (21.8) | 22.4 (20) | 21.5 (18.9) |
60 SPS | Sinc1 | 0.71 (3.9) | 0.043 (0.27) | 0.042 (0.26) | 23.6 (21.1) | 21.8 (19.1) | 20.8 (18.2) |
60 SPS | Sinc2 | 0.6 (3.3) | 0.036 (0.24) | 0.034 (0.21) | 23.8 (21.4) | 22.1 (19.3) | 21.1 (18.5) |
60 SPS | Sinc3 | 0.56 (3) | 0.032 (0.19) | 0.03 (0.17) | 23.9 (21.5) | 22.2 (19.6) | 21.3 (18.8) |
60 SPS | Sinc4 | 0.53 (2.7) | 0.031 (0.19) | 0.03 (0.18) | 24 (21.6) | 22.3 (19.7) | 21.3 (18.7) |
100 SPS | Sinc1 | 0.8 (4.8) | 0.056 (0.34) | 0.054 (0.35) | 23.4 (20.8) | 21.4 (18.8) | 20.5 (17.8) |
100 SPS | Sinc2 | 0.68 (4.2) | 0.047 (0.29) | 0.043 (0.3) | 23.6 (21) | 21.7 (19) | 20.8 (18) |
100 SPS | Sinc3 | 0.67 (4.2) | 0.042 (0.28) | 0.041 (0.27) | 23.6 (21) | 21.8 (19.1) | 20.9 (18.1) |
100 SPS | Sinc4 | 0.62 (3.6) | 0.039 (0.24) | 0.039 (0.27) | 23.8 (21.2) | 21.9 (19.3) | 20.9 (18.2) |
400 SPS | Sinc1 | 1.4 (11) | 0.11 (0.81) | 0.11 (0.75) | 22.6 (19.6) | 20.4 (17.5) | 19.5 (16.7) |
400 SPS | Sinc2 | 1.2 (8.3) | 0.09 (0.64) | 0.086 (0.6) | 22.8 (20) | 20.7 (17.9) | 19.8 (17) |
400 SPS | Sinc3 | 1.1 (7.7) | 0.082 (0.61) | 0.078 (0.56) | 22.9 (20.1) | 20.9 (18) | 19.9 (17.1) |
400 SPS | Sinc4 | 1 (7.7) | 0.076 (0.59) | 0.072 (0.53) | 23 (20.1) | 21 (18) | 20 (17.2) |
1200 SPS | Sinc1 | 2.3 (17) | 0.18 (1.3) | 0.18 (1.4) | 21.9 (19) | 19.7 (16.9) | 18.8 (15.7) |
1200 SPS | Sinc2 | 2 (14) | 0.15 (1.2) | 0.15 (1.1) | 22.1 (19.3) | 20 (17) | 19 (16.1) |
1200 SPS | Sinc3 | 1.8 (13) | 0.14 (1) | 0.13 (1) | 22.2 (19.4) | 20.1 (17.2) | 19.2 (16.2) |
1200 SPS | Sinc4 | 1.7 (13) | 0.13 (1) | 0.13 (0.94) | 22.3 (19.4) | 20.2 (17.2) | 19.2 (16.3) |
2400 SPS | Sinc1 | 3.2 (26) | 0.25 (2) | 0.24 (1.8) | 21.4 (18.4) | 19.2 (16.2) | 18.3 (15.4) |
2400 SPS | Sinc2 | 2.7 (20) | 0.22 (1.7) | 0.21 (1.5) | 21.6 (18.7) | 19.5 (16.5) | 18.5 (15.6) |
2400 SPS | Sinc3 | 2.5 (18) | 0.2 (1.4) | 0.19 (1.4) | 21.7 (18.9) | 19.6 (16.7) | 18.6 (15.8) |
2400 SPS | Sinc4 | 2.3 (18) | 0.18 (1.5) | 0.18 (1.4) | 21.8 (18.9) | 19.7 (16.7) | 18.8 (15.8) |
4800 SPS | Sinc1 | 4.4 (35) | 0.34 (2.5) | 0.32 (2.4) | 20.9 (17.9) | 18.8 (15.9) | 17.9 (15) |
4800 SPS | Sinc2 | 3.9 (30) | 0.3 (2.3) | 0.29 (2.4) | 21.1 (18.1) | 19 (16.1) | 18 (15) |
4800 SPS | Sinc3 | 3.6 (27) | 0.28 (2) | 0.26 (2) | 21.2 (18.3) | 19.1 (16.3) | 18.2 (15.2) |
4800 SPS | Sinc4 | 3.4 (27) | 0.26 (1.9) | 0.25 (1.9) | 21.3 (18.3) | 19.2 (16.3) | 18.2 (15.3) |
7200 SPS | Sinc1 | 5.2 (42) | 0.38 (2.9) | 0.37 (3) | 20.7 (17.7) | 18.6 (15.7) | 17.7 (14.7) |
7200 SPS | Sinc2 | 4.7 (37) | 0.36 (2.8) | 0.34 (2.5) | 20.8 (17.9) | 18.7 (15.8) | 17.8 (14.9) |
7200 SPS | Sinc3 | 4.5 (35) | 0.34 (2.5) | 0.32 (2.4) | 20.9 (18) | 18.8 (15.9) | 17.9 (15) |
7200 SPS | Sinc4 | 4.2 (36) | 0.32 (2.5) | 0.31 (2.3) | 21 (17.9) | 18.9 (16) | 18 (15.1) |
The ADS1235 is a three differential-input, precision 24-bit, ΔΣ ADC with a low-noise PGA and programmable digital filter. The low-noise, low-drift architecture of the PGA makes the ADC suitable for precision measurement of low signal level sensors, such as strain-gauge bridges and resistive pressure transducers. The ADC provides optional chop and ac-bridge excitation modes to eliminate offset drift error.
Key features of the ADC are:
The analog inputs (AINx) connect to the input multiplexer (MUX). The ADC supports three differential or five single-ended input measurement configurations. A second voltage reference input and AC-bridge excitation drive outputs (GPIO) are multiplexed with the analog input pins.
The programmable gain amplifier (PGA) follows the input multiplexer. The gain is programmable to 1, 64 or 128. The PGA bypass option connects the analog inputs directly to the precharge buffered modulator, extending the input voltage range to the voltage of the power supplies. The PGA output connects to pins CAPP and CAPN. The ADC antialias filter is provided at the PGA output with an external capacitor. A monitor is used for detection of PGA overrange conditions.
The delta-sigma modulator measures the differential input voltage relative to the reference voltage to produce the 24-bit conversion result. The differential input range of the ADC is ±VREF / Gain.
The digital filter averages and decimates the modulator output data to yield the final, down-sampled conversion result. The sinc filter is programmable (sinc1 through sinc4) allowing optimization of conversion time, conversion noise and line-cycle rejection. The finite impulse response (FIR) filter mode provides single-cycle settled data with simultaneous rejection of 50-Hz and 60-Hz at data rates of 20 SPS or less.
Two reference voltage input pairs are provided. The primary reference input pair (REFP0/REFN0) is available as standalone input pins. A second reference input pair (REFP1/REFN1) is multiplexed with analog inputs AIN0 and AIN1. A monitor is used for detection of low or missing reference voltage.
The ADC provides four GPIO control lines. The GPIOs are used for input and output of general-purpose logic signals, as well as providing output drive signals for ac-excited bridges. The GPIOs and ac-bridge excitation drive outputs are multiplexed to the analog inputs.
The internal temperature sensor voltage is read by the ADC through the analog input multiplexer.
The SPI-compatible serial interface is used to read the conversion data and also to configure and control the ADC. Data communication errors are detected by CRC. The serial interface consists of four signals: CS, SCLK, DIN and DOUT/DRDY. The dual function DOUT/DRDY provides data output and also the data ready signal. The ADC serial interface can be implemented with as little as three pins by tying CS low.
The ADC clock is either internal or external. The ADC detects the mode of clock operation automatically. The clock frequency is 7.3728 MHz.
Data conversions are controlled by the START pin or by the START command. The ADC is programmable for continuous or one-shot conversions. The DRDY or DOUT/DRDY pin provides the conversion-data ready signal. When taken low, the RESET pin resets the ADC. The ADC is powered down by the PWDN pin or is powered down in software mode.
The ADC operates in either bipolar analog supply configuration (±2.5 V), or in single 5-V supply configuration. The digital power supply range is 2.7 V to 5 V. The BYPASS pin is the internal subregulator output used for the ADC digital core.
The following sections describe the functional blocks of the ADC.
Figure 39 shows the analog input circuit consisting of ESD-protection diodes, the input multiplexer and the PGA. The ADS1235 has six analog inputs to support three differential-input measurement channels. In addition, there are two internal (system) measurements, and an option to disconnect all inputs.
ESD diodes are incorporated to protect the ADC inputs from possible ESD events occurring during the manufacturing process and during PCB assembly when manufactured in an ESD-controlled environment. For system-level ESD protection, consider the use of external ESD protection devices for pins that are exposed to ESD, including the analog inputs.
If either input is driven below AVSS – 0.3 V, or above AVDD + 0.3 V, the internal protection diodes may conduct. If these conditions are possible, use external clamp diodes, series resistors, or both to limit the input current to the specified maximum value.
The input multiplexer selects the signal for measurement. The multiplexer consists of independently programmable positive and negative sections. See Figure 39 for multiplexer register settings. The multiplexers select any input as positive and any input as negative for connection to the PGA. For example, to select AIN5 and AIN4 as a differential input with (+) and (-) polarity, program the INPMUX register to the value of 87h.
When the multiplexer is changed, a break-before-make sequence is performed in order to reduce charge injection into the next measurement channel. Be aware that over-driving unused channels beyond the power supplies can effect conversions taking place on active channels. See the Input Overload section for more information.
The ADC has an internal temperature sensor. The temperature sensor is comprised of two internal diodes with one diode having 80 times the current density of the other. The difference in current density of the diodes yields a differential output voltage that is proportional to absolute temperature. The temperature sensor reading is converted by the ADC. See Figure 39 for register settings to select the temperature sensor for measurement.
Equation 2 shows how to convert the temperature sensor reading to degrees Celsius (˚C):
Measure the temperature sensor with PGA on, gain = 1 and ac-bridge excitation mode disabled. As a result of the low package-to-PCB thermal resistance, the internal temperature closely tracks the PCB temperature.
This configuration opens the inputs to the PGA. Use this configuration to disconnect the PGA from the sensor. With all inputs disconnected, the conversion data are invalid due to the floating input condition. See Figure 39 for the register setting value to open all inputs.
This configuration connects the PGA inputs to the internal VCOM voltage as defined: (AVDD + AVSS) / 2. Use this connection to short the inputs to measure the ADC noise performance and offset voltage, or to short the inputs for offset calibration. See Figure 39 for register settings for the internal VCOM connection.
The analog input pins have multiplexed alternate functions. The alternate functions are the second reference input and GPIO to provide the ac-bridge excitation drive signals. The functions are enabled by programming the associated function registers. The analog inputs retain measurement capability if the alternate functions are programmed. Table 2 summarizes the alternate functions multiplexed to the analog input pins.
ANALOG INPUTS | REFERENCE INPUTS | GPIO/AC-BRIDGE EXCITATION (2-wire mode) | GPIO/AC-BRIDGE EXCITATION (4-wire mode) |
---|---|---|---|
AIN0 | REFP1 | GPIO0/ACX1 | GPIO0/ACX1 |
AIN1 | REFN1 | GPIO1/ACX2 | GPIO1/ACX2 |
AIN2 | GPIO2/ACX1 | ||
AIN3 | GPIO3/ACX2 | ||
AIN4 | |||
AIN5 |
The PGA is a low-noise, CMOS differential-input, differential-output amplifier. The PGA extends the dynamic range of the ADC, important when used with low-level output sensors. Gain is controlled by the GAIN[2:0] register bits as shown in Figure 40. In PGA bypass mode, the input voltage range extends to the analog supplies. The PGA is powered down in bypass mode.
The PGA consists of two chopper-stabilized amplifiers (A1 and A2), and a resistor network that determines the PGA gain. The resistor network is precision-matched, providing low drift performance. The PGA has internal noise filters to reduce sensitivity to electromagnetic-interference (EMI). The PGA output is monitored to provide indication of a possible PGA overload condition.
Pins CAPP and CAPN are the PGA positive and negative outputs, respectively. Connect an external 4.7-nF capacitor (type C0G) as shown in Figure 40. The capacitor filters the sample pulses caused by the modulator, and with the internal resistors the antialias filter is provided. Place the capacitor as close as possible to the pins using short, direct traces. Avoid running clock traces or other digital traces close to these pins.
The input voltage range is determined by the magnitude of the reference voltage and ADC gain. As shown in Figure 19, conversion voltage noise is constant over the specified reference voltage range. Table 3 shows the differential input voltage range verses gain for VREF = 5 V.
GAIN[2:0] BITS | GAIN | FULL-SCALE DIFFERENTIAL INPUT VOLTAGE RANGE(1) |
---|---|---|
000 | 1 | ±5.000 V |
110 | 64 | ±0.078 V |
111 | 128 | ±0.039 V |
As with many amplifiers, the PGA has an input voltage range specification that must not be exceeded in order to maintain linear operation. The input range is specified as an absolute voltage (signal plus common mode voltage) at both positive and negative inputs. As specified in Equation 3, the maximum and minimum absolute input voltage depends on gain, the expected maximum differential voltage, and the minimum value of the analog power supply voltage.
where
The relationship of the PGA input to the PGA output is shown graphically in Figure 41. The PGA output voltages (VOUTP, VOUTN) depend on the respective absolute input voltage, the differential input voltage, and the PGA gain. To maintain the PGA within the linear operating range, the PGA output voltages must be restricted within AVDD – 0.3 V and AVSS + 0.3 V. The diagram depicts a positive differential input voltage that results in a positive differential output voltage.
Bypass the PGA to extend the input voltage range to the analog power supply voltages. In bypass mode, the PGA is bypassed and the analog inputs are connected directly to the precharge buffers of the modulator, thereby extending the input voltage range. Be aware of the increased input current in bypass mode. See the Electrical Characteristics for the input current specification.
The PGA has internal monitors to alarm of possible overrange conditions. Overrange conditions are possible if the signal voltage is over-driven, the common-mode voltage is out of range or if too much gain is used for the normal range of input signal. When overranged, the PGA output nodes are in saturation resulting in invalid conversion data. The high alarm bit asserts high (PGAH_ALM) If either the positive or negative PGA output is greater than AVDD – 0.2 V. Similarly, the low alarm bit asserts high (PGAL_ALM) if either positive or negative PGA output is less than AVSS + 0.2 V. The status of the alarm bits are read in the STATUS byte. The alarm bits are read-only and automatically reset at the start of the next conversion cycle after the overrange condition is cleared. The PGA voltage monitor diagram and threshold values are shown in Figure 42 and Figure 43.
The PGA voltage monitors are fast-responding voltage comparators. Comparator operation is disabled during multiplexer changes to minimize triggering of false alarms. However, it is possible the alarms can trigger on other transient overload conditions that may occur after gain changes, sensor connection changes, and so on.
The ADC requires a reference voltage for operation. The ADC allows two external inputs and the internal analog power supply as reference options. The reference voltage is selected by independent positive and negative multiplexers. The default reference is the 5-V analog power supply (AVDD – AVSS). Figure 44 shows the block diagram of the reference multiplexer.
Program the RMUXP[1:0] and RMUXN[1:0] bits of the REF register to select the positive and negative reference voltages, respectively. The positive reference selections are AVDD, REFP0 and AIN0 (REFP1). The negative reference input selections are internal AVSS, REFN0, AIN1 (REFN1). The reference low-voltage monitor is located after the reference multiplexer. See the Reference Monitor section for more information.
Use the external reference by applying the reference voltage to the designated reference input pins. The reference input pins are differential with positive and negative inputs. Program the reference multiplexer bits RMUXP[1:0] and RMUXN[1:0] to select the respective reference voltage for operation. For example, to select REFP0 and REFN0 as the reference voltage, program the REF register to the value of 0Ah. Follow the specified absolute and differential reference voltage operating conditions, as specified in the Recommended Operating Conditions.
Be aware of the reference input current when reference impedances are present, such as by the use of a resistor divider. Consider the effect of the resistance to system accuracy. Connect a capacitor across the reference input pins to filter noise. When R-C filters are used, match the time constants of the input signal filter to the reference voltage filter to maintain constant conversion noise over the signal operating range.
A third reference option is the 5-V analog power supply (AVDD – AVSS). Select this reference option by programming the REF register to 05h. For 6-wire strain-gauge bridge applications that use excitation-sense connections, or for ac-bridge excitation operation, connect the excitation sense lines to the reference input pins and program the ADC for external reference operation.
The ADC incorporates an internal low-voltage monitor of the reference voltage. As shown in Figure 45 and Figure 46, the REFL_ALM bit of the STATUS byte asserts if the reference voltage (VREF = VREFP – VREFN) falls below 0.4 V. The alarm is read-only and resets at the next conversion after the low reference condition is no longer present.
Use the reference monitor to detect a missing or failed reference voltage. To implement detection of a missing reference, use a 100-kΩ resistor across the reference inputs. If either reference input is disconnected, the resistor biases the differential reference input toward 0 V so that the reference monitor detects the disconnected reference.
The ADC includes four GPIO pins, GPIO0 through GPIO3. The GPIOs are digital inputs/outputs that are referenced to analog AVDD and AVSS. The GPIOs are read and written by the GPIO_DAT bits of the MODE3 register. The GPIOs are multiplexed with analog inputs AIN0 to AIN3. As shown in Figure 47, the GPIOs are configured through a series of programming registers. Bits GPIO_CON[3:0] connect the GPIOs to the associated pin (1 = connect). Bits GPIO_DIR program the direction of the GPIOs; (0 = output, 1 = input). The input voltage threshold is the voltage value between AVDD and AVSS. Bits GPIO_DAT[3:0] are the data values for the GPIOs. Observe that if a GPIO pin is programmed as an output, the value read is the value previously written to the register data, not the actual voltage at the pin.
The GPIOs also provide the ac-bridge excitation drive signals. AC-bridge excitation mode overrides the GPIO register data values. See the AC-Bridge Excitation Mode section for details.
The modulator is an inherently stable, fourth-order, 2 + 2 pipelined ΔΣ modulator. The modulator samples the analog input voltage at a high sample rate (fMOD = fCLK / 8) and converts the analog input to a ones-density bit-stream with the density given by the ratio of the input signal to the reference voltage. The modulator shapes the noise of the converter to high frequency, where the noise is removed by the digital filter.
The ADC operates on the principle of oversampling. Oversampling is defined as the ratio of the sample rate of the modulator to that of the ADC output data rate. Oversampling improves ADC noise by digital bandwidth limiting (low-pass filtering) of the data.
The digital filter receives the modulator output data and produces a high-resolution conversion result. The digital filter low-pass filters and decimates the modulator data (data rate reduction), yielding the final data output. By adjusting the type of filtering, tradeoffs are made between resolution, data throughput and line-cycle rejection.
The digital filter has two selectable modes: sin (x) / x (sinc) mode and finite impulse response (FIR) mode (see Figure 48). The sinc mode provides data rates of 2.5 SPS through 7200 SPS with variable sinc orders of 1 through 4. The FIR filter provides simultaneous rejection of 50-Hz and 60-Hz frequencies with data rates of 2.5 SPS through 20 SPS while providing single-cycle settled conversions.
The sinc filter is comprised of two stages: a fixed-decimation sinc5 filter, followed by a variable-decimation, variable-order sinc filter. The first stage filters and down-samples the input data from the modulator to produce an intermediate data rate of 14400 SPS. The second stage receives the intermediate data to provide final output data rates of 7200 SPS through 2.5 SPS. The second stage has programmable orders of sinc.
The data rate is programmed by the DR[3:0] bits of the MODE0 register. The filter mode is programmed by the FILTER[2:0] bits of the MODE0 register (see Figure 48).
The overall frequency response of the sinc filter is low pass. The filter reduces signal and noise beginning at the -3-dB bandwidth. Changing the data rate and filter order changes the filter bandwidth together with the rate of frequency roll-off. See the Filter Bandwidth section for the bandwidth of the filter settings.
Figure 49 shows the frequency response of the sinc filter at 2400 SPS for various orders of the sinc filter. The peaks and nulls are characteristic of the sinc filter response. The frequency response nulls occur at f (Hz) = N · fDATA, where N = 1, 2, 3 and so on. At the null frequencies, the filter has zero gain. The response nulls are superimposed with the larger nulls beginning at 14400 Hz. The larger nulls are produced by the first stage. The frequency response is similar to that of data rates 2.5 SPS through 7200 SPS. Figure 50 shows the frequency response nulls for 10 SPS.
Figure 51 and Figure 52 show the frequency response of data rates 50 SPS and 60 SPS, respectively. Increase the attenuation at 50 Hz or 60 Hz and harmonics by increasing the order of the sinc filter, as shown in the figures.
Figure 53 and Figure 54 show the detailed frequency response at 50 SPS and 60 SPS, respectively.
The finite impulse response (FIR) filter is a coefficient based filter that provides an overall low-pass filter response. The filter provides simultaneous attenuation of 50 Hz and 60 Hz and harmonics at data rates of 2.5 SPS through 20 SPS. The conversion latency time of the FIR filter data rates is single-cycle. As shown in Figure 48, the FIR filter receives pre-filtered data from the sinc filter. The FIR filter decimates the data to yield the output data rates of 20 SPS. A variable averager (sinc1) provides data rates of 10 SPS, 5 SPS, and 2.5 SPS. Table 4 lists the bandwidth of the data rates in FIR filter mode.
Figure 55 and Figure 56 show the FIR filter attenuation at 50 Hz and 60 Hz provided by a series of response nulls placed close to these frequencies. The response nulls are repeated at harmonics of 50 Hz and 60 Hz.
Figure 57 is the FIR filter response at 10 SPS. As a result of the variable averager used to produce rates of 10 SPS and lower, new frequency nulls are superimposed to the response. The first null appears at the data rate. At 10 SPS, additional nulls occur at frequencies folded around multiples of 20 Hz.
The bandwidth of the digital filter depends on the data rate, filter type and order. Be aware that the bandwidth of the entire system is the combined response of the digital filter, the antialias filter and the use of external analog filters. Table 4 lists the bandwidth of the digital filter versus data rate and filter mode.
DATA RATE (SPS) | -3-dB BANDWIDTH (Hz) | ||||
---|---|---|---|---|---|
FIR | SINC1 | SINC2 | SINC3 | SINC4 | |
2.5 | 1.2 | 1.10 | 0.80 | 0.65 | 0.58 |
5 | 2.4 | 2.23 | 1.60 | 1.33 | 1.15 |
10 | 4.7 | 4.43 | 3.20 | 2.62 | 2.28 |
16.6 | — | 7.38 | 5.33 | 4.37 | 3.80 |
20 | 13 | 8.85 | 6.38 | 5.25 | 4.63 |
50 | — | 22.1 | 16.0 | 13.1 | 11.4 |
60 | — | 26.6 | 19.1 | 15.7 | 13.7 |
100 | — | 44.3 | 31.9 | 26.2 | 22.8 |
400 | — | 177 | 128 | 105 | 91.0 |
1200 | — | 525 | 381 | 314 | 273 |
2400 | — | 1015 | 751 | 623 | 544 |
4800 | — | 1798 | 1421 | 1214 | 1077 |
7200 | — | 2310 | 1972 | 1750 | 1590 |
To reduce 50-Hz and 60-Hz noise interference, configure the data rate and filter to reject noise at 50 Hz and 60 Hz. Table 5 summarizes the 50-Hz and 60-Hz noise rejection versus data rate and filter mode. The table values are based on 2% and 6% tolerance of signal frequency to ADC clock frequency. For the sinc filter, increase noise rejection by increasing the order of the filter. Common-mode noise is also rejected at these frequencies.
DATA RATE (SPS) | FILTER TYPE | DIGITAL FILTER RESPONSE (dB) | |||
---|---|---|---|---|---|
50 Hz ±2% | 60 Hz ±2% | 50 Hz ±6% | 60 Hz ±6% | ||
2.5 | FIR | –113 | –99 | –88 | –80 |
2.5 | Sinc1 | –36 | –37 | –40 | –37 |
2.5 | Sinc2 | –72 | –74 | –80 | –74 |
2.5 | Sinc3 | –108 | –111 | –120 | –111 |
2.5 | Sinc4 | –144 | –148 | –160 | –148 |
5 | FIR | –111 | –95 | –77 | –76 |
5 | Sinc1 | –34 | –34 | –30 | –30 |
5 | Sinc2 | –68 | –68 | –60 | –60 |
5 | Sinc3 | –102 | –102 | –90 | –90 |
5 | Sinc4 | –136 | –136 | –120 | –120 |
10 | FIR | –111 | –94 | –73 | –68 |
10 | Sinc1 | –34 | –34 | –25 | –25 |
10 | Sinc2 | –68 | –68 | –50 | –50 |
10 | Sinc3 | –102 | –102 | –75 | –75 |
10 | Sinc4 | –136 | –136 | –100 | –100 |
16.6 | Sinc1 | –34 | –21 | –24 | –21 |
16.6 | Sinc2 | –68 | –42 | –48 | –42 |
16.6 | Sinc3 | –102 | –63 | –72 | –63 |
16.6 | Sinc4 | –136 | –84 | –96 | –84 |
20 | FIR | –95 | –94 | –66 | –66 |
20 | Sinc1 | –18 | –34 | –18 | –24 |
20 | Sinc2 | –36 | –68 | –36 | –48 |
20 | Sinc3 | –54 | –102 | –54 | –72 |
20 | Sinc4 | –72 | –136 | –72 | –96 |
50 | Sinc1 | –34 | –15 | –24 | –15 |
50 | Sinc2 | –68 | –30 | –48 | –30 |
50 | Sinc3 | –102 | –45 | –72 | –45 |
50 | Sinc4 | –136 | –60 | –96 | –60 |
60 | Sinc1 | –13 | –34 | –12 | –24 |
60 | Sinc2 | –27 | –68 | –24 | –48 |
60 | Sinc3 | –40 | –102 | –36 | –72 |
60 | Sinc4 | –53 | –136 | –48 | –96 |
Conversions are controlled by either the START pin or by the START command. If using commands to control conversions, keep the START pin low to avoid contentions between pin and commands. Commands take affect on the 16th falling SCLK edge (CRC mode disabled) or on the 32nd falling SCLK edge (CRC mode enabled). See Figure 4 for conversion-control timing details.
The ADC provides two conversion modes: continuous and pulse. The continuous-conversion mode performs conversions indefinitely until stopped by the user. Pulse-conversion mode performs one conversion and then stops. The conversion mode is programmed by the CONVRT bit (bit 4 of register MODE0).
This conversion mode performs continuous conversions until stopped by the user. To start conversions, take the START pin high or send the START command. DRDY is driven high at the time the conversion is initiated. DRDY is driven low when the conversion data are ready. Conversion data are available to read at that time. Conversions are stopped by taking the START pin low or by sending the STOP command. When conversions are stopped, the conversion in progress runs to completion. To restart a conversion that is in progress, toggle the START pin low-then-high or send a new START command.
In pulse-conversion mode, the ADC performs one conversion when START is taken high or when the START command is sent. When the conversion completes, further conversions stop. The DRDY output is driven high to indicate the conversion is in progress, and is driven low when the conversion data are ready. Conversion data are available to read at that time. To restart a conversion in progress, toggle the START pin low-then-high or send a new START command. Driving START low or sending the STOP command does not interrupt the current conversion.
The digital filter averages data from the modulator in order to produce the conversion result. The stages of the digital filter must have settled data in order to provide fully-settled output data. The order and the decimation ratio of the digital filter determine the amount of data averaged, and in turn, affect the latency of the conversion data. The FIR and sinc1 filter modes are zero latency because the ADC provides the conversion result in one conversion cycle. Latency time is an important consideration for the data throughput rate in multiplexed applications.
Table 6 lists the conversion latency values of the ADC. Conversion latency is defined as the time from the start of the first conversion, by taking the START pin high or sending the START command, to the time when fully settled conversion data are ready. If the input signal is settled, then the ADC provides fully settled data. The conversion latency values listed in the table are with the start-conversion delay parameter = 50 µs, and include the overhead time needed to process the data. After the first conversion completes (in continuous conversion mode), the period of the following conversions are equal to 1/fDATA. The first conversion latency in chop and ac-excitation modes are twice the values listed in the table. Also when operating in these modes, the period of following conversions are equal to the values listed in the table.
DATA RATE
(SPS) |
CONVERSION LATENCY - t(STDR)(1) (ms) | ||||
---|---|---|---|---|---|
FIR | SINC1 | SINC2 | SINC3 | SINC4 | |
2.5 | 402.2 | 400.4 | 800.4 | 1,200 | 1,600 |
5 | 202.2 | 200.4 | 400.4 | 600.4 | 800.4 |
10 | 102.2 | 100.4 | 200.4 | 300.4 | 400.4 |
16.6 | — | 60.43 | 120.4 | 180.4 | 240.4 |
20 | 52.23 | 50.43 | 100.4 | 150.4 | 200.4 |
50 | — | 20.43 | 40.43 | 60.43 | 80.43 |
60 | — | 17.09 | 33.76 | 50.43 | 67.09 |
100 | — | 10.43 | 20.43 | 30.43 | 40.43 |
400 | — | 2.925 | 5.425 | 7.925 | 10.43 |
1200 | — | 1.258 | 2.091 | 2.925 | 3.758 |
2400 | — | 0.841 | 1.258 | 1.675 | 2.091 |
4800 | — | 0.633 | 0.841 | 1.050 | 1.258 |
7200 | — | 0.564 | 0.702 | 0.841 | 0.980 |
If the input signal changes while free-running conversions, the conversion data are a mix of old and new data, as shown in Figure 58. After an input change, the number of conversion periods required for fully settled data are determined by dividing the conversion latency by the period of the data rate, plus add one conversion period to the result. In chop and ac-bridge excitation modes, use twice the latency values listed in the table.
Some applications require a delay at the start of a conversion in order to allow settling time for the PGA antialias filter or to allow time after input and configuration changes. The ADC provides a user programmable delay time that delays the start of a new conversion. The default value is 50 μs. 50 μs allows for settling of the antialiasing filter placed at the PGA output. Use additional delay time as needed to provide settling time for external components. The delay time increases the conversion latency values listed in Table 6. As an alternative to the programmable start-conversion delay, manually delay the start of conversion after input and configuration changes.
Start-conversion delay is an important consideration for operation in ac-bridge excitation mode. In this mode, the reference inputs to the bridge, and therefore, the bridge output signals are reversed for each conversion. As a result, time delay is required to allow for settling of external filter components after the bridge voltage is reversed. As a general guideline, set the start-conversion delay parameter to a minimum of 15 times the R-C time constant of the signal input and reference input filters.
The PGA and modulator are chopper-stabilized at high frequency in order to reduce offset voltage, offset voltage drift and 1/f noise. The offset and noise artifacts are modulated to a high frequency by the chop operation, which are removed by the digital filter. Although chopper stabilization is designed to remove all offset, a small offset voltage may remain. The optional global chop mode removes the remaining offset errors, providing near zero offset voltage drift performance.
Chop mode alternates the signal polarity between consecutive conversions in order to remove offset. The ADC subtracts consecutive, alternate-polarity conversions to yield the final conversion data. The result of subtraction removes the offset.
As shown in Figure 59, the internal chop switch reverses the signal after the input multiplexer. VOFS models the internal offset voltage. The operational sequence of chop mode is as follows:
Conversion C1: VAINP – VAINN – VOFS → First conversion withheld after start
Conversion C2: VAINN – VAINP – VOFS → Output 1 = (C1 – C2) / 2 = VAINP – VAINN
Conversion C3: VAINP – VAINN – VOFS → Output 2 =-(C3 – C2) / 2 = VAINP – VAINN
The sequence repeats for all conversions. Because of the required settling time to alternate the internal polarity, the effective data rate in chop mode operation is reduced. The chop mode data rate is proportional to the order of the sinc filter. Referring to Table 6, the new data rate is equal to 1 / latency values; and be aware the chop mode first conversion latency is 2 × latency values. As a consequence of the internal data subtraction, two data points are effectively averaged together. Averaging of data reduces noise by √2. Divide the noise data values shown in Table 1 by √2 to derive the chop mode noise performance data. The null frequencies of the digital filter are not changed in chop-mode operation. However, new null frequencies appear at multiples of fDATA / 2 as a result of averaging.
Resistive bridge sensors are excited by dc or ac voltages; or by dc or ac currents. DC voltage excitation is the most common type of excitation. AC excitation reverses the polarity of the excitation voltage by the use of external switching components. Similar in concept to chop mode, the result of the voltage reversal removes offset voltage in the connections leading from the bridge to the ADC inputs. This also includes the offset voltage of the ADC itself. The ADC provides the signals necessary to drive the external switching components in order to reverse the bridge voltage.
The timing of the drive signals is synchronized to the ADC conversion phase. During one conversion phase, the voltage polarity is normal. For the alternate conversion phase, the voltage polarity is reversed. The ADC compensates the reversed polarity conversion by internal reversing the reference voltage. The ADC subtracts the data corresponding to the normal and reverse phases in order to remove offset voltage from the input.
The ADC output drive signals are non-overlapping in order to avoid bridge cross-conduction that can otherwise occur during excitation voltage reversal. The switch rate of the ac-excitation drive signals are performed at the data rate to avoid unnecessary fast switching. See Figure 7 for output drive timing.
Table 7 shows the ac-bridge excitation drive signals and the associated GPIO pins. Program the ac-bridge excitation mode using the CHOP[1:0] bits in register MODE1. AC-bridge excitation can be programmed for two-wire or four-wire drive mode. For two-wire operation, two drive signals are provided on the GPIOs. If needed, use two external inverters to derive four signals to drive discrete transistors. The GPIO drive levels are referred to the 5-V analog supply. Be aware that the ac-bridge excitation mode changes the nominal data rate, depending on the order of the sinc filter. See the Chop Mode section for details of the effective data rate.
DEVICE PIN | GPIO | 2-WIRE MODE (CHOP[1:0] = 10) | 4-WIRE MODE (CHOP[1:0] = 11) |
---|---|---|---|
AIN0 | GPIO0 | ACX1 | ACX1 |
AIN1 | GPIO1 | ACX2 | ACX2 |
AIN2 | GPIO2 | — | ACX1 |
AIN3 | GPIO3 | — | ACX2 |
Operate the ADC with an external clock or with the internal oscillator. The clock frequency is 7.3728 MHz. For external clock operation, apply the clock signal to CLKIN. For internal-clock operation, connect CLKIN to DGND. The internal oscillator begins operation immediately at power-up. The ADC automatically selects the clock mode of operation. Read the clock mode bit in the STATUS register to determine the clock mode.
The ADC has two power-down modes: hardware and software. In both power-down modes, the digital outputs remain driven. The digital inputs must be maintained at VIH or VIL levels (do not float the digital inputs). The internal low-dropout regulator remains on, drawing 25 µA (typical) from DVDD.
Take the PWDN pin low to engage hardware power-down mode. Except for the internal LDO, all ADC functions are disabled. To exit hardware power-down mode (wake-up) take the PWDN pin high. The register values are not reset at wake-up.
Set the PWDN bit (bit 7 of register MODE3) to engage software power-down mode. Similar to the operation of hardware power-down mode, software mode powers down the internal functions except in this case the serial interface. Exit the software power-down mode by clearing the PWDN bit. The register values are not reset.
The ADC is reset in three ways: at power-on, by the RESET pin, and by the RESET command. When reset, the serial interface, conversion-control logic, digital filter, and register values are reset. The RESET bit of the STATUS byte is set to indicate a device reset has occurred by any of the three reset methods. Clear the bit to detect the next device reset. If the START pin is high after reset, the ADC begins conversions.
At power-on, after the supply voltages cross the reset-voltage thresholds, the ADC is reset and 216 fCLK cycles later the ADC is ready for communication. Until this time, DRDY is held low. DRDY is driven high to indicate when the ADC is ready for communication. If the START pin is high, the conversion cycle starts 512 / fCLK cycles after DRDY asserts high. Figure 5 shows the power-on reset behavior.
Reset the ADC by taking the RESET pin low and then returning the pin high. After reset, the conversion starts 512 / fCLK cycles later. See Figure 6 for RESET timing.
Reset the ADC by the RESET command. Toggle CS high to make sure the serial interface resets before sending the command. For applications that tie CS low, see the Serial Interface Auto-Reset section for information on how to reset the serial interface. After reset, the conversion starts 512 / fCLK cycles later. See Figure 6 for timing details.
The ADC incorporates calibration registers and associated commands to calibrate offset and full-scale errors. Calibrate by using calibration commands, or calibrate by writing to the calibration registers directly (user calibration). To calibrate by command, send the offset or full-scale calibration commands. To user calibrate, write values to the calibration registers based on calculations of the conversion data. Perform offset calibration before full-scale calibration.
Use the offset and full-scale (gain) registers to correct offset or full-scale errors, respectively. As shown in Figure 60, the offset calibration register is subtracted from the output data before multiplication by the full-scale register, which is divided by 400000h. After the calibration operation, the final output data are clipped to 24 bits.
Equation 4 shows the internal calibration.
The offset calibration word is 24 bits, consisting of three 8-bit registers, as listed in Table 8. The offset value is subtracted from the conversion result. The offset value is in two's complement format with a maximum positive value equal to 7FFFFFh, and a maximum negative value equal to 800000h. A register value equal to 000000h has no offset correction. Although the offset calibration register provides a wide range of possible offset values, the input signal after calibration cannot exceed ±106% of the pre-calibrated range; otherwise, the ADC is overranged. Table 9 lists example values of the offset register.
REGISTER | BYTE ORDER | ADDRESS | BIT ORDER | |||||||
---|---|---|---|---|---|---|---|---|---|---|
OFCAL0 | LSB | 07h | B7 | B6 | B5 | B4 | B3 | B2 | B1 | B0 (LSB) |
OFCAL1 | MID | 08h | B15 | B14 | B13 | B12 | B11 | B10 | B9 | B8 |
OFCAL2 | MSB | 09h | B23 (MSB) | B22 | B21 | B20 | B19 | B18 | B17 | B16 |
OFCAL[2:0] REGISTER VALUE | IDEAL OUTPUT VALUE(1) |
---|---|
000001h | FFFFFFh |
000000h | 000000h |
FFFFFFh | 000001h |
The full-scale calibration word is 24 bits consisting of three 8-bit registers, as listed in Table 10. The full-scale calibration value is in straight-binary format, normalized to a unity-gain factor at a value of 400000h. Table 11 lists register values for selected gain factors. Gain errors greater than unity are corrected by using full-scale values less than 400000h. Although the full-scale register provides a wide range of possible values, the input signal after calibration must not exceed ±106% of the precalibrated input range; otherwise, the ADC is overranged.
REGISTER | BYTE ORDER | ADDRESS | BIT ORDER | |||||||
---|---|---|---|---|---|---|---|---|---|---|
FSCAL0 | LSB | 0Ah | B7 | B6 | B5 | B4 | B3 | B2 | B1 | B0 (LSB) |
FSCAL1 | MID | 0Bh | B15 | B14 | B13 | B12 | B11 | B10 | B9 | B8 |
FSCAL2 | MSB | 0Ch | B23 (MSB) | B22 | B21 | B20 | B19 | B18 | B17 | B16 |
FSCAL[2:0] REGISTER VALUE | GAIN FACTOR |
---|---|
433333h | 1.05 |
400000h | 1.00 |
3CCCCCh | 0.95 |
The offset self-calibration command corrects offset errors internal to the ADC. When the offset self-calibration command is sent, the ADC disconnects the external inputs, shorts the inputs to the PGA, and then averages 16 conversion results to compute the calibration value. Averaging the data reduces conversion noise to improve calibration accuracy. When calibration is complete, the ADC restores the user input and performs one conversion using the new calibration value.
The offset system-calibration command corrects system offset errors. For this type of calibration, the user shorts the inputs to either the ADC or to the system. When the command is sent, the ADC averages 16 conversion results to compute the calibration value. Averaging the data reduces conversion noise to improve calibration accuracy. When calibration is complete, the ADC performs one conversion using the new calibration value.
The full-scale calibration command corrects gain error. To calibrate, apply a positive full-scale calibration voltage to the ADC, wait for the signal to settle, and then send the calibration command. The ADC averages 16 conversion results to compute the calibration value. Averaging the data reduces conversion noise to improve calibration accuracy. The ADC computes the full-scale calibration value so that the calibration voltage is scaled to positive full scale output code. When calibration is complete, the ADC performs one new conversion using the new calibration value.
Use the following procedure to calibrate using commands. The register-lock mode must be UNLOCK for all calibration commands. After power-on, make sure the reference voltage has stabilized before calibrating. Perform offset calibration before full-scale calibration.
DATA RATE
(SPS) |
FILTER MODE (1) | ||||
---|---|---|---|---|---|
FIR | SINC1 | SINC2 | SINC3 | SINC4 | |
2.5 | 6805 | 6801 | 7601 | 8401 | 9201 |
5 | 3405 | 3401 | 3801 | 4201 | 4601 |
10 | 1705 | 1701 | 1901 | 2101 | 2301 |
16.6 | — | 1021 | 1141 | 1261 | 1381 |
20 | 854.5 | 850.9 | 951.0 | 1051 | 1151 |
50 | — | 340.9 | 380.9 | 420.9 | 460.9 |
60 | — | 284.2 | 317.5 | 350.9 | 384.2 |
100 | — | 170.9 | 190.9 | 210.9 | 230.9 |
400 | — | 43.36 | 48.36 | 53.36 | 58.36 |
1200 | — | 15.02 | 16.69 | 18.36 | 20.02 |
2400 | — | 7.938 | 8.772 | 9.605 | 10.44 |
4800 | — | 4.397 | 4.813 | 5.230 | 5.647 |
7200 | — | 3.216 | 3.494 | 3.772 | 4.050 |