Table 2. Design Parameters

8.2.2.1 Inductor Selection

The TPS61199 is designed to work with inductor values between 10 µH to 47 µH. Running the controller at higher switching frequencies allows the use of smaller and/or lower profile inductors in the 10-µH range. Running the controller at slower switching frequencies requires the use of larger inductors, near 47 µH, to maintain the same inductor current ripple but may improve overall inefficiency due to smaller switching losses. Inductor values can have ±20% tolerance with no current bias. When the inductor current approaches saturation level, its inductance can decrease 20% to 35% from the 0A value depending on how the inductor vendor defines saturation. In a boost regulator, the inductor peak current can be calculated with Equation 6 and Equation 7.

Equation 6.

Equation 7.

where

• V_OUT = output voltage
• I_OUT = total LED current
• V_IN = input voltage
• η = power conversion efficiency, use 85% for TPS61199 applications
• L = inductor value
• F_SW = switching frequency

Select an inductor with a saturation current over the calculated peak current. To calculate the worst case inductor peak current, use the minimum input voltage, maximum output voltage, and maximum total LED current. Select an inductor with a saturation current at least 30% higher the calculated peak current to account for load transients when dimming. Table 3 lists the recommended inductors

8.2.2.2 Schottky Diode

The TPS61199 demands a high-speed rectification for optimum efficiency. Ensure that the average and peak current ratings of the diode exceed the output LED current and inductor peak current. In addition, the reverse breakdown voltage of the diode must exceed the application output voltage. Therefore, TI recommends the VISHAY SS5P9.

8.2.2.3 Switch MOSFET and Gate Driver Resistor

The TPS61199 demands a power N-MOSFET (see Q1 in Typical Application) as a switch. The voltage and current rating of the MOSFET must be higher than the application output voltage and the inductor peak current. The applications benefits from the addition of a resistor (see R8 in Typical Application) connected between the GDRV pin and the gate of the switching MOSFET. With this resistor, the load regulation between LED dimming on and off period and EMI are improved. TI recommends a 3-Ω resistor value. The TPS61199 exhibits lower efficiency when the resistor value is above 3 Ω.

8.2.2.4 Current Sense and Current Sense Filtering

R1 determines the correct overcurrent limit protection. To choose the right value of R1, start with the total system power needed P_OUT. Input current Iin = P_OUT / (V_IN × efficiency). Efficinecy can be estimated from Figure 10. The second step is to calculate the inductor ripple current based on the inductor value L.

Thus, the peak current Ipk = Iin + dIL/2. The maximum R1 can now be calculated as

TI recommends adding 20% or more margin to account for component variations.

A small filter placed on the ISNS pin improves performance of the converter (see R9 and C6 in Typical Application). The time constant of this filter should be approximately 100 ns. The range of R9 must be from about 100 Ω to 1 kΩ for best results. Locate C6 as close as possible to the ISNS pin to provide noise immunity.

8.2.2.5 Output Capacitor

The output capacitor is mainly selected to meet the requirements for output ripple and loop stability of the whole system. This ripple voltage is related to the capacitance of the capacitor and its equivalent series resistance (ESR). Assuming a capacitor with zero ESR, the minimum capacitance needed for a given ripple can be calculated by:

Equation 10.

where

• V_ripplec is the peak to peak output ripple
• D_MAX is the duty cycle of the boost converter.
• D_MAX is equal to approximately (V_{OUT_MAX} – V_{IN_MIN}) / V_{OUT_MAX} in applications.

Care must be taken when evaluating the derating of a capacitor under DC bias. The DC bias can also significantly reduce capacitance. Ceramic device capacitors can loss as much as 50% of its capacitance at its rated voltage. Therefore, leave the margin on the voltage rating to ensure adequate capacitance in the recommendation table.

The ESR impact on the output ripple must be considered as well if tantalum or electrolytic capacitors are used. Assuming there is enough capacitance such that the ripple due to the capacitance can be ignored, the ESR needed to limit the V_ripple is:

Equation 11.

Ripple current flowing through the ESR of a capacitor causes power dissipation in the capacitor. This power dissipation causes a temperature increase internal to the capacitor. Excessive temperature can seriously shorten the expected life of a capacitor. Capacitors have ripple current ratings that are dependent on ambient temperature and must not be exceeded. Therefore, three electrolytic capacitors (UPW2A330MPD6, Nichicon) in parallel reduces the total ESR, shown as in Typical Application.

In a typical application, the output requires a capacitor in the range of 10 µF to 100 µF. The output capacitor affects the small signal control loop stability of the boost converter. If the output capacitor is below the range, the boost regulator may potentially become unstable.

8.2.2.6 Loop Consideration

The COMP pin on the TPS61199 is used for external compensation, allowing the loop response to be optimized for each application. The COMP pin is the output of the internal transconductance amplifier. The external resistor R4, along with ceramic capacitors C4 and C5, are connected to the COMP pin to provide poles and zero. The poles and zero, along with the inherent pole and zero in a peak current mode control boost converter, determine the closed loop frequency response. This is important to converter stability and transient response. For most of the applications, the recommended values of 10 kΩ for R4, 100 nF for C4 and 470 pF for C5 are sufficient. For applications with different components or requirements, see Description Compensating the Current Mode Boost Control Loop for guidance on selecting different compensation components.