EMI occurs when electric or magnetic fields couple and interfere between two or more electronic devices or systems. In an electronic system, voltage ripple can result in conducted noise propagating from one circuit to another, especially when there are shared connections such as power-supply rails.

In a simple example, imagine hearing audible noise in a radio system that goes away when removing or replacing a faulty device. That device could have been generating ripple, causing interference within the audible range and coupling to the audio output.

In a broader sense, EMI is not limited to audible noise and can interfere with power, system inputs, processing and system outputs. International EMI standards such as Comité International Spécial des Perturbations Radioélectriques (CISPR) 25 specify EMI amplitude limits for different frequencies [1]. In Figure 2-1, conducted noise amplitude is represented on the Y axis in decibel microvolts; frequency is represented on the X axis in megahertz. The graph plots CISPR 25 noise-limit lines with peak-level detector limits in red and average levels in blue. Noise detected using a specific test setup and equipment specified by the CISPR standard must remain below these limit lines.

In battery-powered systems (see Figure 3-1), a switched-mode power supply like a buck converter commutates a switch node between a low voltage (GND) and an input voltage. Filtering the switch-node voltage generates an average DC output voltage, which is between the input voltage and GND in the case of a buck converter. The switching causes input ripple at the fundamental switching frequency, and the square edges result in higher-frequency harmonics. Sharper edges with faster slew rates generate higher-frequency harmonic energy. The trapezoidal input current ripple is discontinuous and exhibits the same high-frequency components in the current ripple.

Parasitic capacitances and inductances are nonideal properties of components in the circuit. These parasitics interact negatively with the voltage and current and generate very-high-frequency spikes and ringing noise. On the other hand, jitter and dithering (spread spectrum) can create low-frequency oscillation and noise. Figure 3-2 shows common frequency ranges where noise occurs from these sources.

This document discusses advanced power-converter features that improve upon existing methods to further reduce EMI. The five features entail the use of:

• An active EMI filter (AEF) to improve passive EMI filter performance.
• Dual-random spread spectrum (DRSS) to improve triangular and random spread spectrum.
• True slew-rate control to improve the series boot resistor technique.
• HotRod™ package technology to improve standard wire-bond connections.
• Integrated capacitors to improve external decoupling capacitor performance.

Figure 4-1 illustrates how to mitigate and filter EMI with passive components.

A high-frequency ceramic input capacitor helps supply the power metal-oxide semiconductor field-effect transistors (MOSFETs) with high-frequency energy to improve the switch’s slew-rate and edge characteristics, thus reducing switch ringing and high-frequency noise. Bulk capacitance provides low-frequency damping and prevents resonance between the filter components. A capacitor-inductor-capacitor (CLC) EMI filter, also known as a π filter, filters differential-mode noise from 10 MHz to 100 MHz, depending on the components selected.

Other components include a ferrite bead that provides high differential-mode impedance at very high frequencies. The ferrite bead impedance is typically specified at 100 MHz. A common-mode choke filters common-mode noise that usually occurs at high frequencies (10 MHz to 300 MHz).

Conventional CLC EMI filters use a capacitor and inductor arranged as a low-pass filter to attenuate noise from a noisy node to a quiet node. You can design the CLC components based on the required attenuation. Many tools are available to aid in designing filters, including application notes [2] and spreadsheet tools [3], as shown in Figure 4-2.

Alternatively, you could use Equation 1 to measure or estimate the required attenuation:

where f_SW is the switching frequency, C_IN is the input capacitance, D is the duty cycle, I_DC is the converter-inductor DC current and V_MAX is the decibel microvolt (dBµV) limit at the switching frequency.

Looking again at Figure 4-2 and considering the filter inductor (L_IN) and filter capacitor (C_F), the CLC π filter provides 40 dB of attenuation per decade at frequencies above the cutoff frequency. Figure 3-1 takes the attenuation required at the switching frequency to design the π-filter cutoff frequency:

You can then use Equation 3 to select filter components based on the required cutoff frequency:

After selecting a typical L_IN range between 0.1 µH and 10 µH, you can calculate the required C_F value. Increasing the capacitance value – as opposed to using larger inductance values, since a larger inductance tends to increase the overall filter size – lowers the cutoff frequency. Plus, higher-value inductors tend to have decreased current ratings for a given size. The most space-efficient designs use more parallel capacitors rather than a larger inductor value.

Increasing C_F capacitance with multiple parallel capacitors is usually easier and more space-effective than increasing L_IN to adjust the filter cutoff frequency. Furthermore, a higher inductance will have a decreased current rating for given size.