Hello. My name is Steve Solanyk, and I'm a Product Marketing Engineer in the Lighting Power Products group at Texas Instruments. I'd like to welcome you to this webinar, where I'll be reviewing our latest LED driver offering, the TPS92691 multi-topology LED driver, a device which enables lighting designers the flexibility to optimize their design for a variety of applications.

So why a multi-topology LED driver? Well, LED lighting is found in a variety of end equipment, such as street and area lighting, architectural spaces, factory automation and visual inspection, and in the medical field, such as examination lamps. And there are many other places, as well, where this amazing technology is now being utilized.

But because of the broad range of applications that can utilize this technology, there is no single LED driver topology that will be optimal for every one of them. Some initial LED driver considerations include just the physical arrangement of the LEDs themselves, taking into consideration restrictions with form factor of the end equipment itself; identifying output current and voltage in overall power requirements; and from there, just to begin determining that regulator type.

Other design tradeoffs to be considered include the overall cost of the LED driver solution within the system it'll be placed, the complexity of the design, thermals. Heatsinking is a very important aspect of LED driver design, and should be considered early on in the design process, as it will definitely impact the cost of the system, as well as even the size. Time and design cycle time of the LED driver, and how many times it will need to be refreshed and reused and leveraged. And there are many other things to be considered, as well, when doing a proper LED design.

The TPS92691 is a next-generation DC-to-DC multi-topology LED driver from Texas Instruments, beginning with the LM342x family, which includes the 3421, 23, 24, and 29 devices, which many of the viewers may be familiar with, as they've been found throughout the automotive, industrial, and commercial marketplaces. These are high-side current-sense devices, and offer a range of features, such as fault detection, thermal foldback, as well as analog and PWM dimming capabilities.

The TPS92690 was released a few years ago as a constant current controller tailored for LCD driver topologies that utilize a low-side current-sense architecture. The TPS92691 incorporates many of the best features of its predecessors. A key feature is its 65-volt rail-to-rail current sense capability. This enables it to support driver topologies requiring either a high-side or low-side current sense. Examples include the buck, the boost, buck-boost, SEPIC, and Cuk.

The TPS92691 offers a range of features and benefits for the LED lighting designer. As mentioned before, it does offer a wide input voltage range-- 4 and 1/2 volts to 65 volts. A plus or minus 3% current accuracy is supported by the rail-to-rail current-sense amplifier. And it offers that accuracy up to 140 degrees Celsius. It has dedicated analog and PWM dimming capability. The linear input range for analog dimming is 140 millivolts to 2.5 volts, and over 1,000 to 1 series FET PWM dimming capability with its contrast ratio.

Switching frequency is addressable, along with a synchronization option, thereby allowing for simpler EMI filter design if multiple TPS92691s are utilized in a system. There's also a dedicated IMON pin, which we'll go into further later on in this presentation. But it allows for a continuous LED status-monitoring output. This can be used to help protect against fault in abnormal operating conditions in the LED driver solution. And we do offer an AEC-Q100-qualified version of this device for the automotive and aftermarket automotive marketplaces.

We also have tools and resources to enable you to evaluate the device in a couple of different topologies. We have a SEPIC-based design, and a boost, and boost-to-battery LED driver evaluation board. Along with that, software tools are available, including PSpice. And Texas Instruments' WEBENCH online tool is coming soon.

The TPS92691 is assembled in a 16-lead HTSSOP package. I'll highlight a few of the pins and their associated functions here. But I'll go into more detail later on in this presentation. There is soft-start available on this part, and it is programmable with the SS pin. The RT slash SYNC pin is where the switching frequency is programmed with a resistor to ground. It's also where an external synchronization signal can be applied through a AC-coupled capacitor.

As mentioned before, there are two pins for PWM dimming and for analog dimming. The analog dimming pin is actually labeled the IDJ pin. And it serves as both setting up the LED current reference, but it's also where a voltage can be applied and modulated to implement the analog dimming.

The IMON pin is where the LED current, as measured with a sense resistor between the CSP and CSN pins, is reported out as a voltage. The current as measured by the current-sense amplifiers, translated with some gain into a voltage that is reported out on IMON, and can be used by the LED driver designer to monitor the LED current, and also report out any faults or abnormal operation.

The DDRV is a dedicated gate driver output for implementing series dimming. Overvoltage protection is available through the OVP pin. The TPS92691 uses a peak-current-mode control, and it senses the switch current for that purpose, using the IS pin.

The TPS92691's internal rail-to-rail current-sense amplifier measures the average LED current based on the differential voltage drop across a sense resistor placed between the CSP and CSN inputs. This measurement can be done over a common mode range of zero to 65 volts. Examples of high-side and low-side current sensing are shown with the boost converter examples on the right.

Now while a low-side current-sense option is available with this boost example and its ground reference LED load, for topologies such as the buck-boost, shown in the bottom left, where the LED load is referenced not to ground, but actually back to VN, the flexibilty enabled by the high-side current sense with wide common-mode range becomes quite apparent.

Before moving on, I just wanted to also note that the current-sense amplifier diagram also shows some optional common-mode or differential-mode low-pass filter implementations that can be used to smooth out the effects of large output current ripple and switching current spikes caused by diode reverse recovery. These are detailed further in the product datasheet.

The LED current is set by the IADJ pin voltage and the LED current-sense resistor, RCS. The differential voltage across the sense resistor is regulated to the analog adjust voltage applied to the IADJ pin and scaled by the current-sense amplifier, which has a voltage gain of 14. The LED current can then be adjusted by varying the voltage on the address pin, using either a resistor divider from VCC or an external voltage source. The IADJ pin can also be tied to VCC through an external resistor to set the LED current based on the internal 2.42-volt reference voltage.

A constant-current LCD driver that can be configured for multiple topologies over wide operating conditions is only a benefit to the designer, as long as accurate current regulation can be maintained. The TPS92691 can achieve better than 5% accuracy over a wide range of input voltages and LED stack voltages.

Current accuracy over temperature is also exceptional, as it's maintained with less than 1% error from minus 40 degrees Celsius to over 125 degrees Celsius. This means that [? well-controlled ?] line and load regulation can be expected whether input voltage levels change, the number of LEDs and their configuration requirements change, the LED lighting environment itself changes, or a combination of the situations just mentioned occurs.

Because a dedicated analog adjust capability is available in the form of the IADJ pin, flexible analog LED current adjustment is made available to the designer. In addition to the static reference setting implemented by resistor divider to the VCC pin, the IADJ input can be used in conjunction with an NTC resistor to implement thermal foldback protection. A PWM signal, in conjunction with a first- or second-order low-pass filter, can also be used to program the IADJ voltage. This is a useful option in cases where a microcontroller is also available to the designer to use.

Let's touch on the PWM dimming feature a bit more. As mentioned earlier, the TPS92691 incorporates a dimming input, designated as the PWM pin for pulse-width modulating the output LED current. The brightness of the LEDs can be linearly varied by modulating the duty cycle of the pulsating voltage source connected to that pin.

The DDRV pin's output follows the PWM input signal. And it's capable of sinking and sourcing up to 500 milliamps of peak current to control a low-side series-connected N-channel dimming FET. This series dimming FET is required to achieve high contrast ratio, as it ensures fast rise and fall times with the LED current in response to the PWM input. Without any dimming FET, the rise and fall times are limited by the inductor slew rate and the closed loop bandwidth of the system. This low-side series dimming method is recommended for SEPIC-, Cuk-, and flyback-based LED drivers.

Alternatively, the DDRV output can be translated within an external level-shift circuit to drive a high-side series P-channel dimming FET. As noted in the scope waveform, dimming performance is comparable to the low-side dimming method. And linear and monotonic behavior with over 100 to 1 contrast ratio for dimming frequencies up to 400 hertz is achievable. This low-side series dimming method is recommended for boost-, buck-boost-, and buck-based LCD drivers.

The TPS92691 was designed to ensure a fast transient response with minimum LED current overshoot for a wide dimming range. Driving the PWM input below 2.3 volts turns off switching, [? parks ?] the oscillator, disconnects the COMP pin, and sets the DDRV output to ground in order to maintain the charge on the compensation network and output capacitors.

On the rising edge of the PWM input, when it goes through the 2 and 1/2-volt threshold, the GATE and DDRV outputs are enabled to ramp the inductor current to the previous steady-state value. The COMP pin is connected, and the error amplifier and oscillator are enabled only when the switch current-sense voltage exceeds the COMP voltage, thus immediately forcing the converter into steady-state operation with minimum LED current overshoot.

For low-contrast dimming applications-- say, a dimming ratio of 10 to 1-- PWM dimming can be initiated by just enabling and disabling the PWM input pin, and without a series PWM dimming FET being utilized. This can be useful for cost-sensitive driver designs where high dimming ratios are not required. In this setup, the LED current response and overshoot are controlled by tuning [? them to ?] compensation network.

The soft-start feature helps the regulator gradually reach the steady-state operating point, thus reducing startup stresses and surges. The soft-start time, represented by TSS, is the time required for the LED current to reach the target set point. The required soft-start time is programmed using a capacitor noted as CSS, which is connected from the soft-start pin to ground, and is based on the LED current, output capacitor, and output voltage.

During startup, the TPS92691 clamps the COMP pin to the soft-start pin, which is separated by a diode, until LED current nears the regulation threshold. An internal 10-microamp soft-start current source gradually increases the voltage on the CSS capacitor. This results in a gradual rise of the COMP voltage from ground and ensures the gradual startup.

The TPS92691 switching frequency is programmable by a single external resistor connected between the RT SYNC pin and ground. The relation between switching frequency and RT resistor values is shown in the graph. TI recommends a switching frequency setting between 80 kilohertz and 700 kilohertz for optimal performance over the majority of operating input and output voltage ranges, and for best efficiency.

The internal oscillator can be synchronized by AC coupling an external clock pulse to the RT SYNC pin, as shown in the bottom diagram. The positive-going synchronization clock at the RT pin must exceed the RT SYNC threshold. And the negative-going synchronization clock at the RT pin must exceed the SYNC-falling threshold to trip the internal pulse detector.

TI recommends that the frequency of the external sync pulse is within plus or minus 20% of the internal oscillator frequency programmed by the RT resistor. Additionally, a minimum coupling capacitor of 100 nanofarads and a typical pulse width of 100 nanoseconds should be used for proper synchronization. And in the case where external synchronization is lost, the internal oscillator takes control of the switching rate based on the RT resistor to maintain the output current regulation.

The IMON pin voltage, as mentioned earlier, represents the LED current measured by the rail-to-rail current-sense amplifier across the current-sense resistor. Since the LED current is continuously being measured by the amplifier, the state of the LED current is always being reported on the IMON pin. Under normal operating conditions, the voltage on the IMON pin will equal the voltage on the IADJ pin. However, if the IMON voltage is greater than the IADJ voltage, this would indicate a LED short circuit. And if IMON voltage is less than IADJ voltage, this would indicate a LED open circuit.

And an LCD overcurrent protection example is shown at the bottom. In this particular case, the IMON output can be connected to an external microcontroller or comparator to [? facilitate ?] LED open, short, or cable harness fault detection and mitigation, based on a programmable threshold represented by the symbol VOCTH.

The waveforms here show how the IMON voltage closely tracks the LED current during a LED short-circuit and open-circuit condition, providing immediate fault reporting information. Note, also, that the IMON pin does have an internal clamp of 3.7 volts, which is highlighted in the short-to-ground waveform.

Cycle-by-cycle current limit is accomplished by a redundant internal comparator, which immediately terminates the gate output when the IS input voltage, represented by VIS, exceeds a 525-millivolt limit threshold. Upon a current limit event, the soft-starting COMP pins are internally grounded to reset the state of the controller. The GATE output is enabled after the expiration of the 35-microsecond internal fault timer. And a new startup sequence is initiated through the soft-start pin.

The TPS92691 device includes a dedicated OVP pin, which can be used for either input or output overvoltage protection. This pin features a precision 1.24-volt threshold with 20 microamps of hysteresis current. The overvoltage threshold limit is set by resistor divider network, represented by R OV1 and R OV2, which is connected to the input or output terminal to ground. When the OVP pin voltage exceeds the reference threshold, represented by VOVP THR, the GATE and VDRV pins are immediately pulled low, and the soft-start and COMP capacitors are discharged. The GATE is enabled, and a new startup sequence is initiated after the voltage drops below the hysteresis threshold set by the 20-microamp source current and the external resistor divider.

I'd like to now go over a brief design example using the TPS92691 in a buck-boost topology. Buck-boost LED drivers provide the flexibility needed in applications that support multiple LED load configurations. This means that LED current regulation will occur even when the LED stack voltage goes higher or lower than the input voltage source, or vice versa, where the LED load doesn't change, but the input voltage source can be greater or lower than the output voltage. An example of this could be an application where the LED driver's powered by a battery, but then can also be powered through an off-main AC-to-DC power supply at a higher voltage.

This example is designed to support a wide range of output voltage and LED current specifications. And therefore, it is based on the maximum output power set by the lumen output specified for this application. The design procedure [? is ?] for a battery-connected application with three to nine LEDs in series and a maximum output power of 15 watts. The input voltage range is specced for seven to 18 volts with a typical 14.

A 4 and 1/2-volt UVLO setting, again, with the three to nine LEDs, results in a minimum output voltage of 9.6 volts to a maximum of 28.8 volts. Output current is being specified from 500 milliamps to 1.5 amps. There are LED current ripple specs here of 5%, a dimming range goal of 4% to 100%. And input voltage ripple, OVP requirements, and a soft-start period, along with a target switching frequency.

In regards to the maximum output power, there, again, is the 15-watt requirement. Part of the method involves having a goal of when the output power at the critical conduction mode, the discontinuous conduction mode boundary occurs. And for this design, that's at five watts. Basically, this method of using the maximum output power is to address the wide range of, again, loads, both in LED stack voltage and LED current, with a maximum output power of 15 watts. Meaning the assumption is that, if the voltage of the LED stack increases, the LED current would decrease in order to maintain that 15-watt upper boundary for maximum output power. And vice versa-- if the stack voltage lowers, the LED current can increase, but still remain within that 15-watt boundary condition.

In any case, the additional details of this buck-boost example and the design method based on this maximum output power can be found in the product datasheet. For applications that have, say, a fixed number of LEDs and a narrow LED current range of operation, alternative design equations are also provided in the product datasheet. And those can be used for developing an optimized circuit and build material for that type of design.

To begin, we calculate the typical duty cycle, which, for this example, is a V OUT of 19.2 volts and a V IN of 14 volts. But then we also calculate the maximum and minimum duty cycle, as [? that ?] will come into play later on in other equations. Next, we calculate the resistor RT to set the switching frequency of 390 kilohertz. The inductor is selected to meet the CCM-to-DCM-- or Continuous Conduction Mode to Discontinuous Conduction Mode-- boundary power requirement, which is represented by P sub O BDRY, or what we'll say is just the P OUT boundary.

Typically, the boundary condition is set to enable CCM operation at the lowest possible operating power, based on the minimum LED [? for a ?] voltage drop and LED current. In most applications, this P OUT boundary is set to be about a third of the maximum output power, represented by a P sub O MAX, or P OUT max. The inductor current is calculated for maximum input voltage and output voltage which, in this case, is calculated to be 31 and 1/2 microhenries, roughly. The closest standard inductor value [? of ?] 33 microhenry is then selected, and then [? used ?] itself to calculate the inductor ripple current, which is around 438 milliamps.

This, in turn, is going to be used to determine that we have high enough [? of a ?] current rating for our inductor, so that we don't enter inductor saturation. So the inductor saturation rate should exceed the calculated peak current which, in turn, is based on the maximum output power. And we get that to be about 3.9 amps.

The output capacitor should be selected to meet the 5% peak-to-peak LED current ripple specification. Based on the maximum output power, the capacitance is calculated to be 31 microfarads. Therefore, we could pick a minimum of four 10-microfarad, 50-volt X7R ceramic capacitors placed in parallel to meet this ripple specification. However, additional capacitance may be required based on the derating factor under DC bias operation.

Moving onto the input capacitor, that'll be calculated based on the peak-to-peak input ripple specifications which, for this example, was 70 millivolts over the range of operation. That capacitance is calculated to be about 33 microfarads. Again, a parallel combination of four 10-microfarad, 50-volt X7R ceramic capacitors can be used. And along with the output capacitor, derating should be taken in mind for the input capacitance [? while ?] under DC bias operation.

Next, we calculate the minimum transistor voltage and current rating, which comes out to be roughly 70 volts and 2.8 amps, respectively. There, we could pick a 60-volt or 100-volt N-channel MOSFET with a current rating exceeding three amps.

Moving on, we calculate the minimum Schottky diode voltage and current rating, which comes out to roughly 70 volts and 1 and 1/2 amps. Therefore, we could pick a 60-volt or 100-volt Schottky diode, but with a current rating exceeding 1 and 1/2 amps. One thing to note is that TI typically recommends a single high-current diode instead of parallelling multiple lower current rate diodes to ensure reliable operation over temperature.

Next, we solve for the RIS resistor used to set the switch current limit and slope compensation. Since the TPS92691 utilizes a peak current mode control, slope compensation is needed to eliminate subharmonic oscillations for duty cycles greater than 50%. Two equations are solved for, with the lowest of the two values generated selected for the [? IS sense ?] resistor. In this case, a standard resistor of 0.1 ohms can be picked. And this will ensure stable current-loop operation with none of the previously mentioned subharmonic oscillations occurring over the entire input and output voltage ranges.

Next, we need to program the LED currents per the specified range. This, again, was the 500 milliamps minimum to the 1 and 1/2 amps maximum. The LED current can be programmed to match particular LED string configurations by using a resistor divider, designated with the resistors R ADJ1 and R ADJ2 from VCC to ground for a given sense resistor, RCS.

Regarding RCS, to maximize the accuracy of the driver solution, the IADJ pin voltage is set to 2.1 volts for the specified LED current of 1 and 1/2 amps. That current-sense resistor, RCS, is then calculated as 0.1 ohms, which is a standard resistor value. And along with that, Table 1 then describes the additional R ADJ1 and ADJ2 resistor values that are calculated to program the remaining LED currents.

Compensation is up next. A simple integral compensator provides a good starting point to achieve stable operation across the wide operating range. The product datasheet provides additional guidelines for calculating the compensation required, including the equation shown here. Based on this information, a compensation capacitor of 100 nanofarads will meet our needs.

Next, we calculate the soft-start capacitor needed to meet the specified soft-start period of eight milliseconds. This will be based on the maximum output voltage, minimum LED current. And this results in a capacitance value of around 71 nanofarads. Next, we'll set up the overvoltage protection. Here we calculate the OVP resistor divider values, along with taking into account the OVP hysteresis that's required.

And lastly, for PWM dimming considerations, because we're looking at a buck-boost topology, we'll use a P-channel FET with an external level-shift circuit. The drive strength will will be based on the level-translator resistor and the N-channel offset that's chosen to be driven by the DDRV pin.

And here we have some typical performance curves for this buck-boost example, again, which was based on the maximum output power design method. Very tight line regulation over the range of specified LED stack voltages. Linear analog dimming for analog inputs of 140 millivolts, up to 2.25 volts. And efficiencies that reach, in some cases, up to 90%.

And finally, I'd like to encourage you to visit the TPS92691 product folder, located at the web link listed at the top of this slide. In addition to parametric information and the product datasheet, tools such as our evaluation boards, PSpice model, and user guides are available that can help you evaluate and test out the TPS92691 for your own specific application. Additional support can also be found, including a link to TI's E2E community, where TI has technical staff available to answer any questions you may have regarding the TPS92691 and any other LED lighting needs you may have.