8.1.1.1.2.1 Diode Detector

A diode is one of the simplest types of RF detectors. As depicted in Figure 78, the diode converts the RF input voltage into a rectified current. This unidirectional current charges the capacitor. The RC time constant of the resistor and the capacitor determines the amount of filtering applied to the rectified (detected) signal.

Figure 78. Diode Detector

The advantages and disadvantages can be summarized as follows:

• The temperature stability of the diode detectors is generally very good, since they contain only one semiconductor device that operates at RF frequencies.
• The dynamic range of diode detectors is poor. The conversion gain from the RF input power to the output voltage quickly drops to very low levels when the input power decreases. Typically a dynamic range of 20 dB to 25 dB can be realized with this type of detector.
• The response of diode detectors is waveform dependent. As a consequence of this dependency for example its output voltage for a 0-dBm WCDMA signal is different than for a 0-dBm unmodulated carrier. This is due to the fact that the diode measures peak power instead of average power. The relation between peak power and average power is dependent on the wave shape.
• The transfer shape of diode detectors puts high requirements on the resolution of the ADC that reads their output voltage. Especially at low input power levels a very high ADC resolution is required to achieve sufficient power measurement accuracy (See Figure 76).

8.1.1.1.2.2 (Root) Mean Square Detector

This type of detector is particularly suited for the power measurements of RF modulated signals that exhibits large peak to average power ratio variations. This is because its operation is based on direct determination of the average power and not – like the diode detector – of the peak power.

The advantages and disadvantages can be summarized as follows:

• The temperature stability of (R)MS detectors is almost as good as the temperature stability of the diode detector; only a small part of the circuit operates at RF frequencies, while the rest of the circuit operates at low frequencies.
• The dynamic range of (R)MS detectors is limited. The lower end of the dynamic range is limited by internal device offsets.
• The response of (R)MS detectors is highly waveform independent. This is a key advantage compared to other types of detectors in applications that employ signals with high peak-to-average power variations. For example, the (R)MS detector response to a 0-dBm WCDMA signal and a 0-dBm unmodulated carrier is essentially equal.
• The transfer shape of R(MS) detectors has many similarities with the diode detector and is therefore subject to similar disadvantages with respect to the ADC resolution requirements (see Figure 77).

8.1.1.1.2.3 Logarithmic Detectors

The transfer function of a logarithmic detector has a linear in dB response, which means that the output voltage changes linearly with the RF power in dBm. This is convenient since most communication standards specify transmit power levels in dBm as well.

The advantages and disadvantages can be summarized as follows:

• The temperature stability of the LOG detector transfer function is generally not as good as the stability of diode and R(MS) detectors. This is because a significant part of the circuit operates at RF frequencies.
• The dynamic range of LOG detectors is usually much larger than that of other types of detectors.
• Since LOG detectors perform a kind of peak detection their response is wave form dependent, similar to diode detectors.
• The transfer shape of LOG detectors puts the lowest possible requirements on the ADC resolution (See Figure 77).

8.2.1.2.1.1 Concept of Power Measurements

Power measurement systems generally consists of two clearly distinguishable parts with different functions:

1. A power detector device, that generates a DC output signal (voltage) in response to the power level of the (RF) signal applied to its input.
2. An “estimator” that converts the measured detector output signal into a (digital) numeric value representing the power level of the signal at the detector input.

A sketch of this conceptual configuration is depicted in Figure 80.

Figure 80. Generic Concept of a Power Measurement System

The core of the estimator is usually implemented as a software algorithm, receiving a digitized version of the detector output voltage. Its transfer F_EST from detector output voltage to a numerical output should be equal to the inverse of the detector transfer F_DET from (RF) input power to DC output voltage. If the power measurement system is ideal, that is, if no errors are introduced into the measurement result by the detector or the estimator, the measured power P_EST - the output of the estimator - and the actual input power P_IN should be identical. In that case, the measurement error E, the difference between the two, should be identically zero:

Equation 12.

From the expression above it follows that one would design the F_EST transfer function to be the inverse of the F_DET transfer function.

In practice the power measurement error will not be zero, due to the following effects:

• The detector transfer function is subject to various kinds of random errors that result in uncertainty in the detector output voltage; the detector transfer function is not exactly known.
• The detector transfer function might be too complicated to be implemented in a practical estimator.

The function of the estimator is then to estimate the input power P_IN, that is, to produce an output P_EST such that the power measurement error is - on average - minimized, based on the following information:

1. Measurement of the not completely accurate detector output voltage V_OUT
2. Knowledge about the detector transfer function F_DET, for example the shape of the transfer function, the types of errors present (part-to-part spread, temperature drift) etc.

Obviously the total measurement accuracy can be optimized by minimizing the uncertainty in the detector output signal (select an accurate power detector), and by incorporating as much accurate information about the detector transfer function into the estimator as possible.

The knowledge about the detector transfer function is condensed into a mathematical model for the detector transfer function, consisting of:

• A formula for the detector transfer function.
• Values for the parameters in this formula.

The values for the parameters in the model can be obtained in various ways. They can be based on measurements of the detector transfer function in a precisely controlled environment (parameter extraction). If the parameter values are separately determined for each individual device, errors like part-to-part spread are eliminated from the measurement system.

Errors may occur when the operating conditions of the detector (for example, the temperature) become significantly different from the operating conditions during calibration (for example, room temperature). Examples of simple estimators for power measurements that result in a number of commonly used metrics for the power measurement error are discussed in LOG-Conformance Error, the Temperature Drift Error, the Temperature Compensation and Temperature Drift Error.

8.2.1.2.1.2 RF Input

RF parts typically use a characteristic impedance of 50 Ω. To comply with this standard the LMH2100 has an input impedance of 50 Ω. Using a characteristic impedance other then 50 Ω will cause a shift of the logarithmic intercept with respect to the value given in the 2.7-V DC and AC Electrical Characteristics. This intercept shift can be calculated according to Equation 13.

Equation 13.

The intercept will shift to higher power levels for R_SOURCE > 50 Ω, and will shift to lower power levels for R_SOURCE < 50 Ω.

8.2.1.2.1.3 Output and Reference

The possible filtering techniques that can be applied to reduce ripple in the detector output voltage are discussed in Filtering. In addition two different topologies to connect the LMH2100 to an ADC are elaborated.

8.2.1.2.1.3.1 Filtering

The output voltage of the LMH2100 is a measure for the applied RF signal on the RF input pin. Usually, the applied RF signal contains AM modulation that causes low frequency ripple in the detector output voltage. CDMA signals for instance contain a large amount of amplitude variations. Filtering of the output signal can be used to eliminate this ripple. The filtering can either be realized by a low pass output filter or a low pass feedback filter. Those two techniques are depicted in Figure 81 and Figure 82.

Figure 81. Low Pass Output Filter

Figure 82. Low Pass Feedback Filter

Depending on the system requirements one of the these filtering techniques can be selected. The low pass output filter has the advantage that it preserves the output voltage when the LMH2100 is brought into shutdown. This is elaborated in Output Behavior in Shutdown. In the feedback filter, resistor R_P discharges capacitor C_P in shutdown and therefore changes the output voltage of the device.

A disadvantage of the low pass output filter is that the series resistor R_S limits the output drive capability. This may cause inaccuracies in the voltage read by an ADC when the ADC input impedance is not significantly larger than R_S. In that case, the current flowing through the ADC input induces an error voltage across filter resistor R_S. The low pass feedback filter doesn’t have this disadvantage.

Note that adding an external resistor between OUT and REF reduces the transfer gain (LOG-slope and LOG-intercept) of the device. The internal feedback resistor sets the gain of the transimpedance amplifier.

The filtering of the low pass output filter is realized by resistor R_S and capacitor C_S. The −3 dB bandwidth of this filter can then be calculated by: ƒ_{−3 dB} = 1 / 2πR_SC_S. The bandwidth of the low pass feedback filter is determined by external resistor R_P in parallel with the internal resistor R_TRANS, and external capacitor C_P in parallel with internal capacitor C_TRANS (see Figure 85). The −3 dB bandwidth of the feedback filter can be calculated by ƒ_{−3 dB} = 1 / 2π (R_P//R_TRANS) (C_P + C_TRANS). The bandwidth set by the internal resistor and capacitor (when no external components are connected between OUT and REF) equals ƒ_{−3 dB} = 1 / 2π R_TRANS C_TRANS = 450 kHz.

8.2.1.2.1.4 Interface to the ADC

The LMH2100 can be connected to the ADC with a single-ended or a differential topology. The single ended topology connects the output of the LMH2100 to the input of the ADC and the reference pin is not connected. In a differential topology, both the output and the reference pins of the LMH2100 are connected to the ADC. The topologies are depicted in Figure 83 and Figure 84.

Figure 83. Single-Ended

Figure 84. Differential Application

The differential topology has the advantage that it is compensated for temperature drift of the internal reference voltage. This can be explained by looking at the transimpedance amplifier of the LMH2100 (Figure 85).

Figure 85. Output Stage of the LMH2100

It can be seen that the output of the amplifier is set by the detection current I_DET multiplied by the resistor R_TRANS plus the reference voltage V_REF:

I_DET represents the detector current that is proportional to the RF input power. The equation shows that temperature variations in V_REF are also present in the output V_OUT. In case of a single ended topology the output is the only pin that is connected to the ADC. The ADC voltage for single ended is thus:

A differential topology also connects the reference pin, which is the value of reference voltage V_REF. The ADC reads V_OUT – V_REF:

Equation 16 does not contain the reference voltage V_REF anymore. Temperature variations in this reference voltage are therefore not measured by the ADC.