Figure 2-1 shows the ALM2402F-Q1 under normal quiescent condition being powered from 12 V single supply with its input and output voltage at 6 V mid-supply (no fault condition).

Figure 2-2 shows ALM2402F-Q1 under a fault condition with its output being shorted to an 18 V battery while being powered by a 12 V supply, requiring only protection of a blocking Schottky diode, SD2. Since the absolute maximum rated supply voltage for ALM2402F-Q1 is 18 V, adding the blocking diode, SD2, allows the battery voltage, Vbat, to pull up the amplifier’s positive supply pin, Vsup, to 17.6 V without any damage.

The addition of 1.8kΩ resistor in the feedback, RF, shown in Figure 2-2, limits to 10mA maximum allowable current through the internal back-to-back input protection diodes located between the ALM2402F-Q1 input terminals.

Under the short to battery conditions output of ALM2402F-Q1 sinks 538mA short-circuit current (see Figure 2-2) attempting to bring down the output voltage. At the same time a body diode - an internal drain-to-nwell, p-n junction of P-channel output transistor - supplies 2mA quiescent current to bias the first two internal stages of the amplifier. Under normal operation of the output PMOS transistor, the current flows through the enhanced p-channel from Vcc to Vout. However, under the fault conditions, when Vout is above Vcc, normal operation of PMOS is shut down (Vgs=0) and thus the current flows in a reverse direction from the output, Vout, through drain-to-nwell body diode to a positive supply – see Figure 2-3. For this reason, under fault condition Vsup gets pulled up within a diode drop below Vbat voltage (see Figure 2-2).

Since the absolute maximum rated supply voltage for ALM2402F-Q1 is 18 V, shorting the output above said voltage requires extra circuitry to prevent EOS damage. To this end, Vsup must be protected by Transient Voltage Suppressor (TVS), a fast activated Zener diode with a breakdown voltage of 18 V or lower, from being pulled close to Vbat through the body diode since this would exceed its maximum rated supply voltage (see Figure 3-1). Additionally, the current through the body diode must be limited to a short-circuit level by adding R4 series resistor, which under short to 26 V fault condition results in over 7 V drop across its 7Ω power resistor (total current of ~1.05A)

However, there is a simpler solution to the above problem by using another part - ALM2403-Q1. Because of its higher absolute maximum rated supply voltage, it eliminates the need for extra protection up to 26 V - see Figure 3-2. As was the case with ALM2402F-Q1 shown in Figure 2-2, all that is required here is a blocking diode, SD5, which allows Vsup to be pulled to 25.6 V with no damage.

Nevertheless, as was the case with Vbat above 18 V in case of ALM2402F-Q1, shorting ALM2403-Q1 output above 26 V will similarly require a transient voltage suppressor (TVS) to clamp Vsup at 26 V (or lower). Also, a series output power resistor has to be added to limit the current through internal structures. Therefore, any protection against short to battery or other fault condition must not only protect against over-voltage at the supply pin, Vsup, but also assure that the resulting current through internal and/or external protection structures, does not exceed maximum rated limit specified in the respective product data sheets.

Therefore, for short to battery condition just above 26 V, the simplest way to limit the current through body diode is to include a small resistor inside the feedback loop in series with the ALM2403-Q1 output. For example, under short to 28 V Vbat condition, Isc current (701mA) is being sunk by the amplifier output and adding just 1Ω power resistor, R5, between Vbat and output pin results in safe Vsup of 25.9 V. R5 also limits the body diode current to a safe 713.1mA, thus fully protects ALM2403-Q1 from any damage – see Figure 3-3.

This approach works well for battery voltages up to few volts above the part’s absolute maximum rated supply voltage but requires much higher value series output power resistor for higher Vbat that negatively affects the output voltage swing from the rails, and lowers power efficiency, under normal operating conditions. For example, in order to prevent over-current damage to the part under short to a 48 V battery, one would need to add around 20Ω resistor in series with ALM2403-Q1 output. However, under normal linear conditions, with +/-200mA peak output current, this would lower the output voltage swing to each rail by additional +/-4V (total of 8 V), requiring a significant increase in the minimum supply voltage. This in turn would results in much lower power efficiency of the solution and thus may not be a viable option.

Therefore, in order to protect ALM2403-Q1 under short to high voltage (48 V) Vbat condition, it may be necessary to add an external Schottky diode, SD7, in parallel with the body diode and capable of carrying a very high current – see Figure 3-4. Under such scenario, in order to limit the current through Schottky diode to 20A (or less) this would require adding R66 1Ω power resistor between Vbat and the Schottky diode. Even though for a given current level the forward bias voltage of Schottky diode is much lower than that of the body diode (thus much more of the current would flow through external Schottky and not the body diode), under high current flow of 20A, its forward bias voltage may be 1 V or higher. Therefore, in order to limit the current through the body diode to a safe operating level, it may also be necessary to add another 1Ω series resistor, R6, between SD7 and the amplifier’s output, VF12 – see Figure 3-4.

Most of the ALM2403-Q1 protection circuits discussed so far involved adding a TVS on the supply pin to prevent over-voltage damage and a Schottky diode with a series output power resistor to guard against the over-current damage – these schemes typically work well but are both bulky and costly solutions. However, there is yet another novel approach that greatly limits the current level under fault conditions (thus eliminates the need for the Schottky diode) without limiting the output voltage swing to rail under normal circuit conditions.

Figure 4-1 shows that under a short to 48 V battery fault condition it may take as much as 20Ω of series output power resistor, R7, to limit the current through the body diode to a safe magnitude. This means that under normal operation, with typical current load of +/-200mA, this would result in additional loss of 8 V of the voltage supply headroom, greatly diminishing power efficiency of the solution. However, if the R7 resistor is placed inside the feedback loop and bypassed with a large, non-polarized capacitor, C7, the resolver normal operation performance is hardly affected.

At the typical frequency of 10kHz used in most automotive resolver excitation applications, the impedance of 10uF bypass capacitor is around 1.6Ω, which lowers the output voltage swing to the rail under +/-200mA peak load by just +/-320mV instead of +/-4 V. This means that there is little need to increase supply voltage, which results in higher efficiency of the resolver solution.

The same scheme may easily be employed to protect the ALM2403-Q1 output against a short to a much higher battery voltage, like 100 V, shown in Figure 4-2 above. The only difference between the two fault protections is that it now takes a higher value series power output resistor, R8, to properly limit the current under fault condition. Nevertheless, since the 100Ω series output resistor is similarly bypassed by the same 10uF capacitor, C1, the normal resolver operation of the circuit is hardly effected and thus almost identical as in the case shown in Figure 4-1. It should be noted that the output resistor bypass capacitor’s maximum rated voltage should be sufficient to withstand the DC voltage imposed across it during a fault condition.

The only other consideration of the effects of adding a large series output resistor paralleled with 10uF capacitor inside the feedback loop has to do with their effect on the modified open-loop output impedance, Zo, which controls the stability of the circuit while driving complex loads. However, as may be seen in the simulation of Zo vs frequency graph shown in Figure 4-3, the R||C inclusion inside the feedback loop hardly makes any difference especially at the 10kHz frequency typically used in most automotive resolver applications. Thus, the circuit is capable of driving the same complex loads as in the case with no additional R||C fault protection circuitry.